Rate control for multi-channel communication systems

ABSTRACT

Techniques to determine a set of rates for a set of data streams to be transmitted in a multi-channel communication system. A group of transmission channels to be used for each data stream is initially identified. An equivalent system for each group is then defined to have an AWGN (or flat) channel and a spectral efficiency equal to the average spectral efficiency of the transmission channels in the group. A metric for each group is then derived based on the associated equivalent system, e.g., set to the SNR needed by the equivalent system to support the average spectral efficiency. A rate for each data stream is then determined based on the metric associated with the data stream. The rate is deemed to be supported by the communication system if the SNR required to support the data rate by the communication system is less than or equal to the metric.

BACKGROUND

1. Field

The present invention relates generally to data communication, and morespecifically to techniques for controlling the rate of data transmissionfor multi-channel communication systems.

2. Background

An orthogonal frequency division multiplex (OFDM) communication systemeffectively partitions the overall system bandwidth into multiple(N_(F)) sub-bands, which may also be referred to as frequencysubchannels or frequency bins. Each frequency subchannel is associatedwith a respective subcarrier (or tone) upon which data may be modulated.For an OFDM system, the data to be transmitted (i.e., the informationbits) is first encoded with a particular coding scheme to generate codedbits, and the coded bits are further grouped into multi-bit symbols thatare then mapped to modulation symbols. Each modulation symbolcorresponds to a point in a signal constellation defined by a particularmodulation scheme (e.g., M-PSK or M-QAM) used for data transmission. Ateach time interval that may be dependent on the bandwidth of eachfrequency subchannel, a modulation symbol may be transmitted on each ofthe N_(F) frequency subchannels. OFDM may be used to combat inter-symbolinterference (ISI) caused by frequency selective fading, which ischaracterized by different amounts of attenuation across the systembandwidth.

A multiple-input multiple-output (MIMO) communication system employsmultiple (N_(T)) transmit antennas and multiple (N_(R)) receive antennasfor data transmission. A MIMO channel formed by the N_(T) transmit andN_(R) receive antennas may be decomposed into N_(S) independentchannels, with N_(S)≦min {N_(T), N_(R)}. Each of the N_(S) independentchannels may also be referred to as a spatial subchannel of the MIMOchannel and corresponds to a dimension. The MIMO system can provideimproved performance (e.g., increased transmission capacity) if theadditional dimensionalities created by the multiple transmit and receiveantennas are utilized.

For a MIMO system that employs OFDM (i.e., a MIMO-OFDM system), N_(F)frequency subchannels are available on each of the N_(S) spatialsubchannels for data transmission. Each frequency subchannel of eachspatial subchannel may be referred to as a transmission channel.N_(F)·N_(S) transmission channels are thus available for datatransmission between the N_(T) transmit antennas and N_(R) receiveantennas.

For a MIMO-OFDM system, the N_(F) frequency subchannels of each spatialsubchannel may experience different channel conditions (e.g., differentfading and multipath effects) and may achieve differentsignal-to-noise-and-interference ratios (SNRs). Each transmittedmodulation symbol is affected by the response of the transmissionchannel via which the symbol was transmitted. Depending on the multipathprofile of the communication channel between the transmitter andreceiver, the frequency response may vary widely throughout the systembandwidth for each spatial subchannel, and may further vary widely amongthe spatial subchannels.

For a multipath channel with a frequency response that is not flat, theinformation rate (i.e., the number of information bits per modulationsymbol) that may be reliably transmitted on each transmission channelmay be different from transmission channel to transmission channel. Ifthe modulation symbols for a particular data packet are transmitted overmultiple transmission channels, and if the response of thesetransmission channels varies widely, then these modulation symbols maybe received with a wide range of SNRs. The SNR would then varycorrespondingly across the entire received packet, which may then makeit difficult to determine the proper rate for the data packet.

Since different receivers may experience different (and possibly widelyvarying) channel conditions, it would be impractical to transmit data atthe same transmit power and/or data rate to all receivers. Fixing thesetransmission parameters would likely result in a waste of transmitpower, the use of sub-optimal data rates for some receivers, andunreliable communication for some other receivers, all of which leads toan undesirable decrease in system capacity. Moreover, the channelconditions may vary over time. As a result, the supported data rates forthe transmission channels would also vary over time. The differenttransmission capabilities of the communication channels for differentreceivers plus the multipath and time-variant nature of thesecommunication channels make it challenging to efficiently transmit datain a MIMO-OFDM system.

There is therefore a need in the art for techniques to control the rateof data transmission in multi-channel communication systems such asMIMO-OFDM systems.

SUMMARY

Techniques are provided herein to control the rate of data transmissionin a multi-channel communication system having multiple transmissionchannels. In an aspect, the rate of each data stream is determined basedon a metric associated with the data stream. This metric may be derivedbased on an equivalent system that models the group of transmissionchannels to be used for the data stream. The equivalent system isdefined to have an AWGN channel (i.e., a flat frequency response) and aspectral efficiency S_(equiv) that is equal to the average spectralefficiency S_(avg) of the group of transmission channels (i.e., theequivalent system has a total capacity equal to the total capacity ofthe group of transmission channels).

A specific embodiment provides a method for determining a set of ratesfor a set of data streams to be transmitted over a wirelesscommunication channel in a multi-channel communication system (e.g., aMIMO-OFDM system). In the method, a group of transmission channels to beused for each data stream is initially identified.

An equivalent system for each transmission channel group is then definedbased on one or more estimated channel characteristics of thetransmission channels in the group. In an embodiment, the equivalentsystem for each transmission channel group may be defined by (1)obtaining an estimate of the SNR of each transmission channel, (2)estimating the spectral efficiency of each transmission channel based onthe estimated SNR and a spectral efficiency function, ƒ(x), and (3)determining the average spectral efficiency of the transmission channelsin the group based on the estimated spectral efficiencies of theindividual transmission channels. The equivalent system is defined tohave an AWGN channel and a spectral efficiency equal to the averagespectral efficiency of the group of transmission channels.

A metric for each transmission channel group is then derived based onthe associated equivalent system. In an embodiment, the metric is set tothe SNR needed by the equivalent system to support the average spectralefficiency. This SNR is referred to as the equivalent SNR and may bedetermined based on an inverse function ƒ⁻¹(x).

A rate for each data stream is then determined based on the metricassociated with the data stream. This may be achieved by evaluating oneor more available rates. For each evaluated rate, the SNR required tosupport the data rate by the communication system is determined, andthis rate is deemed to be supported by the communication system if therequired SNR is less than or equal to the metric.

Various aspects and embodiments of the invention are described infurther detail below. The invention further provides methods, receiverunits, transmitter units, receiver systems, transmitter systems,systems, and other apparatuses and elements that implement variousaspects, embodiments, and features of the invention, as described infurther detail below.

BRIEF DESCRIPTION OF THE DRAWINGS

The features, nature, and advantages of the present invention willbecome more apparent from the detailed description set forth below whentaken in conjunction with the drawings in which like referencecharacters identify correspondingly throughout and wherein:

FIG. 1A is a diagram of a model of a multi-channel communication system;

FIG. 1B is a diagram that graphically illustrates rate selection for amulti-channel communication system with multipath channel based on anequivalent system;

FIG. 2 is a flow diagram of an embodiment of a process for determiningthe maximum data rate supported by a SISO-OFDM system based on anequivalent system;

FIG. 3 is a diagram illustrating the spectral efficiency of theSISO-OFDM system with a multipath channel;

FIG. 4A shows a plot of required SNRs versus data rates for a systemthat supports a set of discrete data rates;

FIG. 4B graphically illustrates the determination of the amount of backoff to use when evaluating whether or not a particular data rate issupported;

FIG. 5A is a diagram illustrating the spectral efficiencies of thespatial subchannels in a MIMO-OFDM system with a multipath channel;

FIG. 5B is a diagram illustrating the spectral efficiency of anequivalent SISO system used to model the MIMO-OFDM system shown in FIG.5A;

FIG. 6 is a flow diagram of an embodiment of a process for controllingthe rate of one or more independently processed data streams in amulti-channel system;

FIG. 7 is a block diagram of an embodiment of a transmitter system and areceiver system in the multi-channel system;

FIG. 8 is a block diagram of a transmitter unit in the transmittersystem; and

FIGS. 9 and 10 are block diagrams of two embodiments of a receiverprocessor in the receiver system.

DETAILED DESCRIPTION

An orthogonal frequency division multiplex (OFDM) communication systemeffectively partitions the overall system bandwidth into multiple(N_(F)) sub-bands, which may also be referred to as frequencysubchannels or frequency bins. Each frequency subchannel is associatedwith a respective subcarrier (or tone) upon which data may be modulated.

A multiple-input multiple-output (MIMO) communication system employsmultiple (N_(T)) transmit antennas and multiple (N_(R)) receive antennasfor data transmission, and is denoted as an (N_(T), N_(R)) system. AMIMO channel formed by the N_(T) transmit and N_(R) receive antennas maybe decomposed into N_(S) independent channels, with N_(S)≦min {N_(T),N_(R)}. Each of the N_(S) independent channels may also be referred toas a spatial subchannel of the MIMO channel. The number of spatialsubchannels is determined by the number of eigenmodes for the MIMOchannel, which in turn is dependent on a channel response matrix, H(k),that describes the response between the N_(T) transmit and N_(R) receiveantennas. For simplicity, in the following description, the channelresponse matrix, H(k), is assumed to be full rank and the number ofspatial subchannels is given as N_(S)=N_(T)≦N_(R).

The rate control techniques described herein may be used for variousmulti-channel communication systems having multiple transmissionchannels that may be used for data transmission. Such multi-channelsystems include MIMO systems, OFDM systems, MIMO-OFDM systems, and soon. The transmission channels may be (1) spatial subchannels in MIMOsystems, (2) frequency subchannels in OFDM systems, or (3) frequencysubchannels of spatial subchannels in MIMO-OFDM systems.

FIG. 1A is a diagram of a model of a multi-channel communication system100. At a transmitter 110, traffic data is provided from a data source112 to a transmit (TX) data processor 114. TX data processor 114 maydemultiplex the traffic data into N_(D) data streams, N_(D) is anyinteger one or greater. Each data stream may be independently processedand then transmitted over a respective group of transmission channels.For each data stream, TX data processor 114 codes the data in accordancewith a particular coding scheme, interleaves the coded data inaccordance with a particular interleaving scheme, and modulates theinterleaved data in accordance with a particular modulation scheme. Themodulation (i.e., symbol mapping) may be achieved by grouping sets ofcoded and interleaved bits to form multi-bit symbols and mapping eachmulti-bit symbol to a point in a signal constellation corresponding tothe selected modulation scheme (e.g., QPSK, M-PSK, or M-QAM). Eachmapped signal point corresponds to a modulation symbol.

In an embodiment, for each data stream, the data rate is determined by adata rate control, the coding scheme is determined by a coding control,and the modulation scheme is determined by a modulation control. Thecontrols are provided by a controller 130 based on feedback informationreceived from a receiver 150.

A pilot may also be transmitted to the receiver to assist it perform anumber of functions such as channel estimation, acquisition, frequencyand timing synchronization, coherent data demodulation, and so on. Inthis case, pilot data is provided to TX data processor 114, which thenprocesses and multiplexes the pilot data with the traffic data.

For OFDM, within a transmitter (TMTR) 116, the modulated data (i.e., themodulation symbols) to be transmitted from each transmit antenna istransformed to the time domain by an inverse fast Fourier transform(IFFT) unit to provide OFDM symbols. Each OFDM symbol is a timerepresentation of a vector of N_(F) modulation symbols to be transmittedon N_(F) frequency subchannels of one transmit antenna in a transmissionsymbol period. In contrast to a single carrier “time-coded” system, anOFDM system effectively transmits the modulation symbols “in thefrequency domain”, by sending in the time domain the IFFT of themodulation symbols for the traffic data.

Transmitter 116 provides an OFDM symbol stream for each transmit antennaused for data transmission. Each OFDM symbol stream is further processed(not shown in FIG. 1A for simplicity) to generate a correspondingmodulated signal. Each modulated signal is then transmitted from arespective transmit antenna over a wireless communication channel to thereceiver. The communication channel distorts the modulated signals witha particular channel response and further degrades the modulated signalswith additive white Gaussian noise (AWGN) having a variance of N₀.

At receiver 150, the transmitted modulated signals are received by eachreceive antenna, and the received signals from all receivers areprovided to a receiver (RCVR) 160. Within receiver 160, each receivedsignal is conditioned and digitized to provide a corresponding stream ofsamples. For each sample stream, a fast Fourier transformer (FFT)receives and transforms the samples to the frequency domain to provide acorresponding received symbol stream. The received symbol streams arethen provided to a receive (RX) data processor 162.

RX data processor 162 processes the received symbol streams to providedecoded data for the transmitted data streams. The processing by RX dataprocessor 162 may include spatial or space-time processing, demodulation(i.e., symbol demapping), deinterleaving, and decoding. RX dataprocessor 162 may further provide the status of each received datapacket. Channel estimator 164 processes the “detected” symbols fromdemodulator/decoder 162 to provide estimates of one or morecharacteristics of the communication channel, such as the channelfrequency response, the channel noise variance N₀ thesignal-to-noise-and-interference ratio (SNR) of the detected symbols,and so on. Typically, only the pilot symbols are used to obtainestimates of the SNR. However, the SNR may also be estimated based ondata symbols, or a combination of pilot and data symbols, and this iswithin the scope of the invention.

A rate selector 166 receives the channel estimates from channelestimator 164 and possibly other parameters and determines a suitable“rate” for each data stream. The rate is indicative of a set ofparameter values to be used for subsequent transmission of the datastream. For example, the rate may indicate (or may be associated with) aspecific data rate to be used for the data stream, a specific codingscheme and/or coding rate, a specific modulation scheme, and so on.

A controller 170 receives the rate(s) from rate selector 166 and thepacket status from RX data processor 162 and provides the appropriatefeedback information to transmitter 110. This feedback information mayinclude the rate(s), the channel estimates, some other information, orany combination thereof. The feedback information may be used toincrease the efficiency of the system by adjusting the processing at thetransmitter such that data is transmitted at the best known settings ofpower and rates supported by the communication channel. The feedbackinformation is then sent back to transmitter 110 and used to adjust theprocessing of the data transmission to receiver 150. For example,transmitter 110 may adjust the data rate, the coding scheme, themodulation scheme, or any combination of the above (based on thefeedback information) for each data stream to be transmitted to receiver150.

In the embodiment shown in FIG. 1A, the rate selection is performed byreceiver 150 and the selected rate for each data stream is provided totransmitter 110. In other embodiments, the rate selection may beperformed by the transmitter based on feedback information provided bythe receiver, or may be performed jointly by both the transmitter andreceiver.

In a single-carrier communication system, the transmitted symbols mayall be received at similar SNR at the receiver. The relationship betweenthe SNR of a “constant SNR” data packet and the probability of error(PE) for the packet is well known in the art. As an approximation, themaximum data rate supported by the single-carrier system with aparticular SNR may be estimated as the maximum data rate supported by anAWGN channel with the same SNR. The main characteristic of the AWGNchannel is that its frequency response is flat or constant across theentire system bandwidth.

However, in a multi-channel communication system, the modulation symbolsthat make up a data packet may be transmitted across multiple frequencysubchannels and/or multiple spatial subchannels. Typically, thecommunication channel between the transmitter and receiver is not flat,but is instead dispersive or frequency selective, with different amountsof attenuation at different sub-bands of the system bandwidth. Moreover,for a MIMO channel, the frequency response for each spatial subchannelmay be different from that of the other spatial subchannels. Thus,depending on the characteristics of the transmission channels used totransmit the packet, the SNR may vary across the entire packet. Thisproblem of “varying SNR” packet is exacerbated for wider systembandwidth and for a multipath channel. For the multipath channel, thedata rate to use for each data stream may be selected to account for themultipath or frequency selective nature of the communication channel.

A major challenge for a multi-channel communication system is then todetermine the maximum data rate that may be used for each data streamwhile achieving a particular level of performance, which may bequantified by a particular packet error rate (PER), frame error rate(FER), block error rate (BLER), bit error rate (BER), or any othercriterion that may be used to quantify performance. For example, thedesired level of performance may be achieved by maintaining the PERwithin a small window around a particular nominal value (e.g.,P_(e)=1%).

Techniques are provided herein to control the rate of data transmissionin a multi-channel communication system with multipath channel. In anaspect, the rate of each data stream is determined based on a metricassociated with the data stream. This metric may be derived based on anequivalent system that models the group of transmission channels usedfor the data stream, as described in further detail below.

FIG. 1B is a diagram that graphically illustrates rate selection for amulti-channel communication system with multipath channel based on anequivalent system. For a given multipath channel defined by a channelresponse of h(k) and a noise variance of N₀, a theoretical multi-channelsystem may be capable of supporting a spectral efficiency of S_(avg)using modulation scheme M, where M may be different for differentfrequency subchannels. As used herein, spectral efficiency representsthe general concept of “capacity per dimension”, where the dimension maybe frequency and/or space. Spectral efficiency is normally given inunits of bits per second per Hertz (bps/Hz). As used herein, atheoretical system is one without any losses, and a practical system isone with (1) implementation losses, e.g., due to hardware imperfections,and (2) code loss due to the fact that practical codes do not work atcapacity. This S_(avg) relates to the average spectral efficiency of thetheoretical system given the channel conditions h(k) and N₀. The averagespectral efficiency S_(avg) may be determined based on a spectralefficiency function ƒ(x), where x denotes a set of input parameters forthe function ƒ(.), as described below.

An equivalent system with an AWGN channel is able to support thespectral efficiency of S_(avg) with an SNR of SNR_(equiv). Thisequivalent system is also a theoretical system. The equivalent SNR,SNR_(equiv), may be derived for spectral efficiency of S_(avg) usingmodulation scheme M and based on a function g(x)=ƒ⁻¹(x), where ƒ⁻¹(x) isan inverse function of ƒ(x).

A practical multi-channel system with an AWGN channel is able to supportdata rate R using modulation scheme M and coding scheme C for a PER ofP_(e) with an SNR of SNR_(req). This data rate R is normalized tobits/sec/Hertz, which is the same unit used for spectral efficiency. Therequired SNR, SNR_(req), may be determined based on computer simulation,empirical measurement, or some other means, and may be stored in atable. The function of required SNR versus data rate is dependent on thespecific modulation scheme M and coding scheme C selected for use. Adata rate is deemed to be supported by the practical multi-channelsystem with multipath channel if the required SNR for the data rate isless than the equivalent SNR. As data rate R increases, the required SNRincreases for the practical system while the equivalent SNR isapproximately constant (except for the variation due to a dependency onmodulation scheme M) since it is defined by the channel conditions h(k)and N₀. The maximum data rate that may be supported by the practicalmulti-channel system with the multipath channel is thus limited by thechannel conditions.

For clarity, the rate control is first described for a single-inputsingle-output (SISO) system, then expanded to cover a single-inputmultiple-output (SIMO) system, and then finally to a MIMO system. In thefollowing description, the SISO, SIMO, and MIMO systems all employ OFDM.

SISO System

For the SISO-OFDM system, there is only one spatial subchannel and thechannel response is defined by {h(k)}, for k=0, 1, . . . (N_(F)−1). Fora multipath channel with channel response of {h(k)} and noise varianceof N₀, these parameters may be mapped to an SNR(k) for each frequencysubchannel k. If the total transmit power, P_(total), for the SISO-OFDMsystem is fixed and the allocation of the transmit power to the N_(F)frequency subchannels is uniform and fixed, then the SNR of eachfrequency subchannel k may be expressed as: $\begin{matrix}{{{SNR}(k)} = {\frac{P_{total}}{N_{F}}{\frac{{{h(k)}}^{2}}{N_{0}}.}}} & {{Eq}\quad(1)}\end{matrix}$

The spectral efficiency of each frequency subchannel k with SNR(k) maybe estimated based on the function ƒ(x), which may be a constrained orunconstrained spectral efficiency function. The absolute orunconstrained spectral efficiency of a system is typically given as thetheoretical maximum data rate that may be reliably transmitted over achannel with a given channel response and noise variance. Theconstrained spectral efficiency of a system is further dependent on thespecific modulation scheme or signal constellation used for datatransmission. The constrained spectral efficiency (due to the fact thatthe modulation symbols are restricted to specific points on the signalconstellation) is lower than the absolute spectral efficiency (which isnot confined by any signal constellation).

In one embodiment, the function ƒ(x) may be defined based on theconstrained spectral efficiency function ƒ_(const)(k), which may beexpressed as: $\begin{matrix}\begin{matrix}{{f_{const}(k)} = {M_{k} - {\frac{1}{2^{M_{k}}}{\sum\limits_{i = 1}^{2^{M_{k}}}{E\left\lbrack {\log_{2}{\sum\limits_{j = 1}^{2^{M_{k}}}{\exp\left( {{- {{SNR}(k)}}\left( {{{a_{i} - a_{j}}}^{2} +}\quad \right.}\quad \right.}}}\quad \right.}}}}} \\{\quad{\left. \quad\left. \quad\left. \quad{2\quad{Re}\left\{ {\beta^{*}\left( {a_{i} - a_{j}} \right)} \right\}} \right) \right) \right\rbrack,}}\end{matrix} & {{Eq}\quad(2)}\end{matrix}$where

-   -   M_(k) is related to modulation scheme M(k), i.e., modulation        scheme M(k) corresponds to a 2^(M) ^(k) -ary constellation        (e.g., 2^(M) ^(k) -ary QAM), where each of the 2^(M) ^(k) points        in the constellation may be identified by M_(k) bits;    -   a_(i) and a_(j) are the points in the 2^(M) ^(k) -ary        constellation;    -   β is a complex Gaussian random variable with zero mean and a        variance of 1/SNR(k); and    -   E[.] is the expectation operation, which is taken with respect        to the variable β in equation (2).        Equation (2) shows that a different modulation scheme M(k) may        be used for each frequency subchannel. For simplicity, one        modulation scheme M may be used for all N_(F) frequency        subchannels for data rate R (i.e., M (k)=M for all k).

The constrained spectral efficiency function ƒ_(const)(k) shown inequation (2) does not have a closed form solution. Thus, this functionmay be numerically derived for various modulation schemes and SNRvalues, and the results may be stored in one or more tables. Thereafter,the function ƒ_(const)(k) may be evaluated by accessing the proper tablewith a specific modulation scheme and SNR.

In another embodiment, the function ƒ(x) is defined based on the Shannon(or theoretical) spectral efficiency function ƒ_(unconst)(k), which maybe expressed as:ƒ_(unconst)(k)=log₂[1+SNR(k)]  Eq (3)As shown in equation (3), the Shannon spectral efficiency is notconstrained by any given modulation scheme (i.e., M(k) is not aparameter in equation (3)).

The spectral efficiency functions provide the spectral efficiency of asystem based on the set of input parameters. These spectral efficiencyfunctions are related to channel capacity functions, which provide the(constrained or unconstrained) capacity of a channel. Spectralefficiency (which is typically given in units of bps/Hz) is related tocapacity (which is typically given in bps) and may be viewed as beingequal to normalized capacity.

The particular choice of function to use for ƒ(x) may be dependent onvarious factors. For a typical system that employs one or more specificmodulation schemes, it has been found that the use of the constrainedspectral efficiency function ƒ_(const)(k) for the function ƒ(x) resultsin accurate estimation of the maximum data rate supported by theSISO-OFDM system with multipath channel.

In a typical communication system, a set of discrete data rates,R={R(r), r=1, 2, . . . P}, may be defined, and only these data rates maybe available for use. Each data rate R(r) in set R may be associatedwith a specific modulation scheme or signal constellation M(r) and aspecific coding rate C(r). Each data rate would further require an SNRof SNR_(req)(r) or better to achieve the desired PER of P_(e). ThisSNR_(req)(r) is determined for the practical SISO-OFDM system with anAWGN channel.

Each data rate R(r) may thus be associated with a set of parameters thatcharacterizes it. These parameters may include the modulation schemeM(r), the coding rate C(r), and the required SNR_(req)(r), as follows:R(r)⇄[M(r), C(r), SNR_(req)(r)]  Eq (4)where r is an index for the data rates, i.e., r=1, 2, . . . , P, and Pis the total number of data rates available for use. Expression (4)states that data rate R(r) may be transmitted using modulation scheme M(r) and coding rate C(r) and further requires SNR_(req)(r) to achievethe desired PER of P_(e).

FIG. 2 is a flow diagram of an embodiment of a process 200 fordetermining the maximum data rate supported by the SISO-OFDM systembased on an equivalent system. For this embodiment, the constrainedspectral efficiency function shown in equation (2) is used for ƒ(x) todetermine the average spectral efficiency of the transmission channelsused for data transmission. Since each data rate R(r) may be associatedwith a different modulation scheme M(r), and since the constrainedspectral efficiency function is dependent on M(r), the average spectralefficiency of the transmission channel may be different for differentdata rates. The equivalent system is dependent on the average spectralefficiency and is thus determined for each data rate in FIG. 2.

Initially, the P data rates supported by the SISO-OFDM system may beordered such that R(1)<R(2)< . . . <R(P). The highest available datarate R(P) is then selected (e.g., by setting the variable r to the indexfor the highest data rate, i.e., r=P) (step 212). Parameter valuesassociated with (1) the transmission channels used for datatransmission, such as the channel response h(k) and the noise varianceN₀, and (2) the selected data rate R(r), such as the modulation schemeM(r), are then determined (step 214). Depending on the design of theSISO-OFDM system, each data rate may be associated with one or multiplemodulation schemes. For simplicity, the following assumes that only onemodulation scheme is associated with each data rate.

The average spectral efficiency S_(avg) of the transmission channels isthen determined (step 216). This may be achieved by first determiningthe SNR(k) of each transmission channel, as shown above in equation (1).Using the constrained spectral efficiency function, the spectralefficiency of each transmission channel is then estimated for SNR(k) andmodulation scheme M(r), as shown in equation (2). The spectralefficiencies of the N_(F) frequency subchannels are then averaged toobtain the average spectral efficiency S_(avg), as follows:$\begin{matrix}{S_{avg} = {\frac{\sum\limits_{k = 0}^{N_{F} - 1}{f(x)}}{N_{F}}.}} & {{Eq}\quad(5)}\end{matrix}$

FIG. 3 is a diagram illustrating the spectral efficiency of theSISO-OFDM system with the multipath channel. For a multipath channelwith varying SNR across the system bandwidth, the SISO-OFDM system isassociated with different spectral efficiencies for different frequencysubchannels, as shown by plot 310. The spectral efficiencies of allN_(F) frequency subchannels used for data transmission may be averagedto obtain the average spectral efficiency S_(avg), which is shown byplot 312. The average spectral efficiency, S_(avg), may be viewed as thespectral efficiency for each of the N_(F) frequency subchannels in theSISO-OFDM system if the communication channel is an AWGN channel insteadof a multipath channel. The constrained or unconstrained spectralefficiency function may thus be used to map a multipath channel to anequivalent AWGN channel.

Referring back to FIG. 2, a metric Ψ is then determined based on anequivalent system (step 218). The equivalent system is defined to havean AWGN channel and an average spectral efficiency S_(equiv) that isequal to the average spectral efficiency of the SISO-OFDM system withthe multipath channel (i.e., S_(equix)=S_(avg)) The SNR needed by theequivalent system to support a data rate of S_(equiv) may then bedetermined based on the inverse of the function used to derive theS_(avg), which in this case is the constrained spectral efficiencyfunction. The metric Ψ may then be set equal to the equivalent SNR, asfollows:Ψ=g(x)=ƒ⁻¹(x)  Eq (6)where ƒ⁻¹(x) denotes the inverse function of ƒ(x). The metric Ψ and theequivalent SNR are both indicative of the “goodness” of the N_(F)frequency subchannels.

The constrained spectral efficiency function ƒ(x) takes two inputs,SNR(k) and M(r), and maps them to a spectral efficiency value. Theinverse constrained spectral efficiency function ƒ⁻¹(x) takes twoinputs, S_(avg) and M(r), and maps them to an SNR value. The functiong(S_(avg), M(r)) thus determines the SNR needed in the equivalent systemto support a spectral efficiency equal to the average spectralefficiency S_(avg) given that constellation M(r) is used. The metric Ψmay thus be determined once for each modulation scheme (i.e., eachsignal constellation). The function g(x) may also be determined forvarious modulation schemes and stored in a table.

The required SNR, SNR_(req)(r), needed to transmit the selected datarate R(r) at the desired PER of P_(e) by the practical SISO-OFDM systemis then determined (step 220). The required SNR is a function of themodulation scheme M(r) and coding rate C(r) associated with the selecteddata rate R(r). The required SNR may be determined for each of thepossible data rates by computer simulation, empirical measurements, orby some other means, and may be stored in a table for later use.

A determination is then made whether or not the selected data rate R(r)is supported by the SISO-OFDM system (step 222). This may be achieved bycomparing the metric Ψ against the required SNR determined for theselected data rate. If the metric Ψ is greater than or equal to therequired SNR (i.e., Ψ≧SNR_(req)(r)), which indicates that the SNRachieved by the SISO-OFDM system for the multipath channel is sufficientto support data rate R(r) for the desired PER of P_(e), then that datarate is selected for use (step 226). Otherwise, the next lower availabledata rate is selected for evaluation (e.g., by decrementing the variabler, or r=r−1) (step 224). This next lower data rate is then evaluated byreturning to step 214. Steps 214 through 222 may be repeated as often asneeded until either (1) the maximum supported data rate is identifiedand provided in step 226 or (2) all available data rates have beenevaluated.

The metric Ψ is dependent on the channel conditions (e.g., h(k) and N₀)and the modulation scheme M(r) if the constrained spectral efficiencyfunction is used. The required SNR is a monotonic function thatincreases with increasing data rate. The embodiment shown in FIG. 2evaluates the available data rates, one at a time, from the maximumavailable data rate to the minimum available data rate. The highest datarate associated with a required SNR that is less than or equal to themetric Ψ is selected for use.

The metric Ψ may be determined based on equations (2), (5), and (6). Inequation (5), a summation is performed for ƒ(x) to accumulate thespectral efficiencies of the individual frequency subchannels to providethe spectral efficiency for the N_(F) frequency subchannels. The averagespectral efficiency S_(avg) is then obtained by dividing the spectralefficiency for the N_(F) frequency subchannels by the number offrequency subchannels. The function g(S_(avg), M(r)) then determines theequivalent SNR for the equivalent system needed to reliably transmitdata at a spectral efficiency equal to the average spectral efficiencyS_(avg) using modulation scheme M(r).

Equation (5) assumes that the same modulation scheme M(r) is used forall N_(F) frequency subchannels in the SISO-OFDM system. Thisrestriction can simplify the processing at the transmitter and receiverin the system but may sacrifice performance.

The metric Ψ may also be defined for the case in which differentmodulation schemes are used for different frequency subchannels. The useof different modulation schemes and/or coding rates for differentfrequency subchannels is often referred to as “bit loading”.

In FIG. 2, the equivalent system is determined for each data rate beingevaluated. This implementation covers a scheme whereby different datarates may be associated with different modulation schemes. However, ifdifferent data rates are associated with the same modulation scheme,then the equivalent system only needs to be determined for eachdifferent modulation scheme that may be used with the data rates beingevaluated. This would then simplify the computation.

As a further simplification, if the average spectral efficiency S_(avg)of the frequency subchannels is only dependent on SNR(k) and not on themodulation scheme, which would be the case if the unconstrained spectralefficiency function is used for ƒ(x), then the equivalent system onlyneeds to be evaluated once, instead of for each data rate evaluated. Theequivalent SNR for the equivalent system can be determined once in themanner described above. Thereafter, the required SNR for each data rate(starting with the highest data rate) may be compared against theequivalent SNR.

In an alternative embodiment, the metric Ψ is defined as apost-detection SNR achieved for the multipath channel by asingle-carrier communication system after equalization. Thepost-detection SNR is representative of the ratio of the total signalpower to the noise plus interference after equalization at the receiver.Theoretical values of post-detection SNR achieved in the single-carriersystem with equalization may be indicative of the performance of aSISO-OFDM system, and therefore may be used to determine the maximumsupported data rate in the SISO-OFDM system. Various types of equalizermay be used to process the received signal in the single-carrier systemto compensate for distortions in the received signal introduced by themultipath channel. Such equalizers may include, for example, a minimummean square error linear equalizer (MMSE-LE), a decision feedbackequalizer (DFE), and others.

The post-detection SNR for an (infinite-length) MMSE-LE may be expressedas: $\begin{matrix}{{{SNR}_{{mmse}\text{-}{le}} = \frac{1 - J_{\min}}{J_{\min}}},} & {{Eq}\quad\left( {7a} \right)}\end{matrix}$where J_(min) is given by $\begin{matrix}{{J_{\min} = {\frac{T}{2\quad\pi}{\int_{{- \pi}/T}^{\pi/T}{\frac{N_{0}}{{X\left( {\mathbb{e}}^{j\quad\omega\quad T} \right)} + N_{0}}{\mathbb{d}\omega}}}}},} & {{Eq}\quad\left( {7b} \right)}\end{matrix}$where X(e^(jωT)) is the folded spectrum of the channel transfer functionH(f).

The post-detection SNR for an (infinite-length) DFE may be expressed as:$\begin{matrix}{{SNR}_{dfe} = {{\exp\left\lbrack {\frac{T}{2\quad\pi}{\int_{{- \pi}/T}^{\pi/T}{{\ln\left( \frac{{X\left( {\mathbb{e}}^{j\quad\omega\quad T} \right)} + N_{0}}{N_{0}} \right)}{\mathbb{d}\omega}}}} \right\rbrack} - 1.}} & {{Eq}\quad(8)}\end{matrix}$The post-detection SNRs for the MMSE-LE and DFE shown in equations (7)and (8), respectively, represent theoretical values. The post-detectionSNRs for the MMSE-LE and DFE are also described in further detail by J.G. Proakis, in a book entitled “Digital Communications”, 3rd Edition,1995, McGraw Hill, sections 10-2-2 and 10-3-2, respectively, which areincorporated herein by reference.

The post-detection SNRs for the MMSE-LE and DFE may also be estimated atthe receiver based on the received signal, as described in U.S. patentapplication Ser. Nos. 09/826,481 and 09/956,449, both entitled “Methodand Apparatus for Utilizing Channel State Information in a WirelessCommunication System,” respectively filed Mar. 23, 2001 and Sep. 18,2001, and U.S. patent application Ser. No. 09/854,235, entitled “Methodand Apparatus for Processing Data in a Multiple-Input Multiple-Output(MIMO) Communication System Utilizing Channel State Information,” filedMay 11, 2001, all assigned to the assignee of the present applicationand incorporated herein by reference.

Post-detection SNRs, such as those described by the analyticalexpressions shown in equations (7) and (8), may be determined for themultipath channel and used as an estimate of the metric Ψ (i.e.,Ψ≈SNR_(mmse-le) or Ψ≈SNR_(dfe)). The post-detection SNR (e.g.,SNR_(mmse-le) or SNR_(dfe)) for the equivalent AWGN channel may becompared against the required SNR, SNR_(req)(r), derived for aparticular set of parameter values, R(r), M(r), C(r), and P_(e), todetermine the data rate that may be used in the SISO-OFDM system withmultipath channel.

The equivalent system that models the transmission channels used for adata stream may be defined to have an AWGN channel and a spectralefficiency equal to the average spectral efficiency of the transmissionchannels. The equivalent system may also be defined based on thepost-detection SNR achieved for the multipath channel by asingle-carrier communication system. The equivalent system may also bedefined in some other manner, and this is within the scope of theinvention. The metric Ψ may also be defined based on some otherfunctions and/or in some other manner, and this is within the scope ofthe invention.

The data rate selected for use in the SISO-OFDM system using the metricΨ represents a prediction of the data rate that may be supported by themultipath channel for the desired PER of P_(e). As with any rateprediction scheme, there will inevitably be prediction errors. In orderto ensure that the desired PER can be achieved, the prediction errorsmay be estimated and a back-off factor may be used in determining thedata rate that can be supported by the multipath channel. This back offreduces the throughput of the system. Thus, it is desirable to keep thisback off as small as possible while still achieving the desired PER. Anaccurate prediction scheme like the one described herein will minimizethe back off to be applied and hence maximize the system capacity.

FIG. 4A shows a plot of required SNRs versus data rates for a systemthat supports a set of discrete data rates. In FIG. 4A, the discretedata rates are labeled as R(r), for r=1, 2, . . . P, on the horizontalaxis. Each data rate R(r) is associated with a respective SNR requiredto achieve the desired PER of P_(e) for the system with an AWGN channel.The required SNRs are labeled as SNR_(req)(r) on the vertical axis. Thediscrete operating points at (R(r), SNR_(req)(r)), for r=1, 2, . . . P,correspond to the minimum SNRs required to support the correspondingdata rates, and are shown by the solid circles 412. A spectralefficiency function for this system is represented by plot 410 (thethick solid line).

For a given multipath channel, the average spectral efficiency S_(avg)may be determined as shown in equation (5), and the metric Ψ for thisaverage spectral efficiency may be determined as shown in equation (6).Graphically, Ψ and S_(avg) may be represented by a point 414 in FIG. 4A,which is marked with an “x”. If this point is in the shaded region aboveplot 410, then the selected data rate associated with Ψ and S_(avg) isdeemed to be supported by the system.

It may be necessary to back off the selected data rate since it is basedon a theoretical value. For example, code loss and implementation lossesmay result in a higher required SNR to achieve the desired PER.Implementation losses due to imperfections in the receiver's pre-decoderstages will show up in the SNR, and losses due to imperfections in thedecoder and the transmitter are typically negligible. The amount of codeloss versus capacity may be estimated and accounted for with a back off.The amount of back off to be used to account for code loss may bedetermined as described below.

FIG. 4B graphically illustrates the determination of the amount of backoff to use when evaluating whether or not a particular data rate issupported. As described above, the set {SNR_(req)(r)}, for r=1, 2, . . .P, represents the SNR required in a practical system to obtain thedesired PER of P_(e). An ideal SNR may be determined for each data ratebased on the (constrained or unconstrained) spectral efficiency functionand is shown on the right vertical axis. The set {SNR_(cap)(r)}, forr=1, 2, . . . P, represents the SNR required in an ideal system (i.e.,with no implementation losses) to obtain the desired PER of P_(e). Itcan be noted that SNR_(cap)(r)<SNR_(req)(r) for all r, sinceSNR_(cap)(r) is the required SNR for an ideal system while SNR_(req)(r)is the required SNR for a practical system. A set {ΔSNR(r)}, for r=1, 2,. . . P, may be defined to represent the additional SNR required for thepractical system to account for losses in the practical system (whichmainly include code loss).

The average spectral efficiency S_(avg) determined in equation (5) willlie between two consecutive data rates, e.g., R(r) and R(r+1), whichhave been normalized to bits/sec/Hertz. The corresponding back offs inSNR at these two data rates are ΔSNR(r) and ΔSNR(r+1), respectively. Inan embodiment, the amount of back off to use for the metric Ψ may bedetermined by linear interpolation of ΔSNR(r) and ΔSNR(r+1), as follows:$\begin{matrix}{{\Delta\quad\Psi} = {\frac{{\Delta\quad{{{SNR}(r)}\left\lbrack {{R\left( {r + 1} \right)} - C_{avg}} \right\rbrack}} + {\Delta\quad{{{SNR}\left( {r + 1} \right)}\left\lbrack {C_{avg} - {R(r)}} \right\rbrack}}}{{R\left( {r + 1} \right)} - {R(r)}}.}} & {{Eq}\quad(9)}\end{matrix}$A backed-off metric, Ψ_(BO), may then be expressed as:Ψ_(BO)=Ψ−ΔΨ  Eq (10)

Referring back to FIG. 2, the backed-off metric, Ψ_(BO), (instead of themetric Ψ) may be compared against the required SNR in step 222 todetermine whether or not the selected data rate R(r) is supported by theSISO-OFDM system.

SIMO System

For a SIMO system, N_(R) receive antennas are used to receive a datatransmission from a single transmit antenna. The channel responsebetween the single transmit antenna and the N_(R) receive antennas maybe represented as h(k) or {h_(i)(k)} for i=1, 2, . . . N_(R) and k=0, 1,. . . (N_(F)−1), where h_(i)(k) is the coupling (i.e., the complex gain)between the transmit antenna and the i-th receive antenna for the k-thfrequency subchannel.

The spectral efficiency function for a (1, N_(R)) SIMO system is thesame as that for a SISO system, except that the received SNR for theSIMO system is obtained by a summation of all the received SNRs for theN_(R) received antennas. Thus, the received SNR for the k-th frequencysubchannel in a SIMO-OFDM system may be expressed as: $\begin{matrix}{{{{SNR}(k)} = \frac{\sum\limits_{i = 1}^{N_{R}}{{h_{i}(k)}}^{2}}{N_{0}}},} & {{Eq}\quad(11)}\end{matrix}$where the transmit power for each frequency subchannel is normalizedto 1. For simplicity, equation (11) assumes that the same noise varianceN₀ is received on all N_(R) receive antennas. Equation (11) may bemodified to account for different noise variances N₀ being received fordifferent receive antennas. In comparison, the received SNR for the k-thfrequency subchannel in a SISO-OFDM system may be expressed as shown inequation (1). For the SIMO-OFDM system, the received SNR determined inequation (11) may then be used in the spectral efficiency function ƒ(x).Except for the change in the computation of SNR, the rate control forthe SIMO-OFDM system may be performed in similar manner as describedabove for the SISO-OFDM system.

MIMO System

For a MIMO-OFDM system, the response between the N_(T) transmit andN_(R) receive antennas may be described by an N_(R)×N_(T) channelimpulse response matrix, H. The elements of the matrix H are composed ofchannel impulse vectors {h _(i,j)}, for i=1, 2, . . . N_(R) and j=1, 2,. . . N_(T), where h _(i,j) describes the coupling between the j-thtransmit antenna and the i-th receive antenna. Each vector h _(i,j) iscomposed of L taps and may be expressed as:h _(i,j) =[h _(i,j)(1)h _(i,j)(2) . . . h _(i,j)(L)]^(T)  Eq (12)where each of the L taps may be modeled as a complex Gaussiancoefficient for a Rayleigh fading channel. For a given (i,j)transmit-receive antenna pair, the signal transmitted from the j-thtransmit antenna may be received by the i-th receive antenna via anumber of propagation paths, and the multipath components associatedwith these propagation paths are assumed to be uncorrelated. This may beexpressed as:E[h _(i,j)(p)h* _(i,j)(q)]=E[|h _(i,j)(p)|²]δ_(p-q)  Eq (13)where p and q represent two multipath components, h* is the complexconjugate of h, and δ_(p-q) is the Delta-Dirac function that is equal toone only if p=q and equal to zero otherwise. Furthermore, it is assumedthat the channel responses for different transmit-receive antenna pairsare uncorrelated, i.e.,${{E\left\lbrack {{\underset{\_}{h}}_{m,n}{\underset{\_}{h}}_{i,j}^{H}} \right\rbrack} = 0},$for different values of m, n, i, and j, where h ^(H) represents theconjugate transpose of h.

The channel impulse response matrix, H(n), is a time-domainrepresentation of the MIMO channel response. A corresponding channelfrequency response matrix, H(k), may be obtained by performing a fastFourier transform (FFT) on H(n), which may be expressed as:H (k)=FFT[H (n)]  Eq (14)where k=0, 1, . . . (N_(F)−1) and N_(F)≧L. In particular, an N_(F)-pointFFT may be performed on a sequence of N_(F) sampled values for a givenelement h _(i,j) of H to derive a sequence of N_(F) coefficients for thecorresponding element h _(i,j) of H. Each element of H is thus the FFTof a corresponding element of H. Each element of H is a vector of N_(F)complex values (i.e., h _(i,j)=[h_(i,j)(0) h_(i,j)(1) . . .h_(i,j)(N_(F)−1)]^(T)), which are representative of the frequencyresponse of the propagation path for a particular (i,j) transmit-receiveantenna pair. The matrix H may thus be viewed as comprising a sequenceof N_(F) matrixes H(k), for k=0, 1, . . . (N_(F)−1), each of dimensionN_(R)×N_(T).

For a MIMO-OFDM system, data may be processed and transmitted usingnumerous processing schemes. Each processing scheme may designate (1)the manner in which data is processed (i.e., encoded, interleaved, andmodulated) prior to transmission, and (2) the transmission channels usedto transmit each independently processed data stream.

In an all antenna processing (AAP) scheme, one data stream istransmitted over all transmit antennas and frequency subchannels. Forthis scheme, the data to be transmitted may be encoded, interleaved,modulated, and then demultiplexed into N_(T) symbol streams for theN_(T) transmit antennas. For the AAP scheme, a coded data packet may beinterleaved in both the frequency and space domains.

In a per-antenna processing (PAP) scheme, one data stream is transmittedover all frequency subchannels of each transmit antenna. For thisscheme, the data to be transmitted is first demultiplexed to N_(T) datastreams for the N_(T) transmit antennas. Each data stream isindependently coded, interleaved, modulated, and then transmitted overone of the N_(T) transmit antennas. The data rates and the coding andmodulation schemes may be the same or different for the N_(T) datastreams. For the PAP scheme, each data stream is only interleaved in thefrequency domain.

Each independently processed data stream may comprise one or more codeddata packets or codewords. Each such codeword is generated at thetransmitter by encoding a packet of data based on a particular codingscheme, and may be decoded at the receiver based on a complementarydecoding scheme. The decoding of each codeword may be achieved by firstrecovering the modulation symbols transmitted for that codeword. Theprocessing scheme selected for use at the transmitter affects theprocessing schemes available for use at the receiver.

The model for the MIMO-OFDM system may be expressed as:y (k)= H (k) x (k)+ n , for k=0, 1, . . . (N _(F)−1),  Eq (15)where y(k) is a vector of N_(R) received symbols for the k-th frequencysubchannel (i.e.,

-   -   the “received” vector for tone k), which may be represented as        y(k)=[y₁(k) y₂(k) . . . y_(N) _(R) (k)]^(T), where y_(i)(k) is        the entry received by the i-th receive antenna for tone k and        i=1,2, . . . , N_(R);    -   x(k) is a vector of N_(T) modulation symbols for tone k (i.e.,        the “transmitted” vector), which may be represented as        x(k)=[x₁(k) x₂(k) . . . x_(N) _(T) (k)]^(T), where x_(j)(k) is        the modulation symbol transmitted from the j-th transmit antenna        for tone k and j=1, 2, . . . , N_(T);    -   H(k) is the channel frequency response matrix for the MIMO        channel for tone k; and    -   n is the additive white Gaussian noise (AWGN) with a mean vector        of 0 and a covariance matrix of Λ _(n)=N₀ I, where 0 is a vector        of zeros, I is the identity matrix with ones along the diagonal        and zeros everywhere else, and N₀ is the noise variance.        For simplicity, the effects of the OFDM processing at both the        transmitter and receiver (which may be negligible) are not shown        in equation (15).

Due to scattering in the propagation environment, the N_(T) symbolstreams transmitted from the N_(T) transmit antennas interfere with eachother at the receiver. In particular, a given symbol stream transmittedfrom one transmit antenna may be received by all N_(R) receive antennasat different amplitudes and phases. Each received symbol stream may theninclude a component of each of the N_(T) transmitted symbol streams. TheN_(R) received symbol streams would collectively include all N_(T)transmitted symbols streams. However, these N_(T) symbol streams aredispersed among the N_(R) received symbol streams.

At the receiver, various processing techniques may be used to processthe N_(R) received symbol streams to detect the N_(T) transmitted symbolstreams. These receiver processing techniques may be grouped into twoprimary categories:

-   -   spatial and space-time receiver processing techniques (which are        also referred to as equalization techniques), and    -   “successive nulling/equalization and interference cancellation”        receiver processing technique (which is also referred to as        “successive interference cancellation” (SIC) processing        technique).        The spatial and space-time receiver processing techniques may        provide better performance for the AAP scheme, while the SIC        processing technique may provide better performance for the PAP        scheme. These receiver processing techniques are described in        further detail below.

For clarity, the following terminology is used herein:

-   -   “transmitted” symbol streams—the modulation symbol streams        transmitted from the transmit antennas;    -   “received” symbol streams—the inputs to a spatial or space-time        processor (in the first stage of a SIC receiver, if one is used,        as shown in FIG. 10);    -   “modified” symbol streams—the inputs to the spatial or        space-time processor in each subsequent stage of the SIC        receiver;    -   “detected” symbol streams—the outputs from the spatial or        space-time processor (up to N_(T)−l+1 symbol streams may be        detected at stage l for a SIC receiver); and    -   “recovered” symbol stream—a symbol stream that is recovered at        the receiver to obtain a decoded data stream (only one detected        symbol stream is recovered at each stage of a SIC receiver).

The spatial and space-time receiver processing techniques attempt toseparate out the transmitted symbol streams at the receiver. Eachtransmitted symbol stream may be “detected” by (1) combining the variouscomponents of the transmitted symbol stream in the N_(R) received symbolstreams based on an estimate of the channel response and (2) removing(or canceling) the interference due to the other transmitted symbolstreams. Each receiver processing technique attempts to either (1)decorrelate the individual transmitted symbol streams such that there isno interference from the other transmitted symbol streams or (2)maximize the SNR of each detected symbol stream in the presence of noiseand interference from the other symbol streams. Each detected symbolstream is then further processed (e.g., demodulated, deinterleaved, anddecoded) to obtain the corresponding data stream.

For simplicity, it is assumed that a linear zero-forcing (ZF) equalizerperforms spatial processing by projecting the received symbol streamsover an interference-free sub-space to obtain the transmitted symbolstreams. The linear ZF equalizer has a response W _(ZF)(k), which may beexpressed as:

  W _(ZF)(k)= H( k)( H ^(H)(k)H(k))⁻¹  Eq (16)

The detected symbol streams,

, which are estimates of the transmitted symbol streams, x, may beestimated as: $\begin{matrix}{{\hat{\underset{\_}{x}}(k)} = {{{{\underset{\_}{W}}_{ZF}^{H}(k)}{\underset{\_}{y}(k)}} = {{\underset{\_}{x}(k)} + {{{\underset{\_}{W}}_{ZF}^{H}(k)}{\underset{\_}{n}.}}}}} & {{Eq}\quad(17)}\end{matrix}$As shown in the right-hand side of equation (17), the detected symbolstreams,

, comprise the transmitted symbol streams, x, plus filtered noise,${{{\underset{\_}{W}}_{ZF}^{H}(k)}\underset{\_}{n}},$which is in general correlated with a covariance matrix$\Sigma_{n} = {N_{0}{\underset{\_}{W}}_{ZF}^{H}{{\underset{\_}{W}}_{ZF}.}}$The correlation takes place on the same frequency subchannel between thedifferent transmit antennas. This correlation is thus applicable forsystems that use all antenna processing (AAP).

The analysis may also be performed based on other linear receivers, asis known by one skilled in the art.

The successive interference cancellation receiver processing techniqueattempts to recover the transmitted symbol streams, one stream at eachstage, using spatial or space-time receiver processing. As each symbolstream is recovered, the interference caused by the recovered symbolstream on the remaining not yet recovered symbol streams is estimatedand canceled from the received symbol streams, and the modified symbolstreams are similarly processed by the next stage to recover the nexttransmitted symbol stream.

For a SIC receiver, the l-th stage initially performs spatial orspace-time processing on the N_(R) modified symbol streams to attempt toseparate out the (N_(T)−l+1) transmitted symbol streams that have notyet been recovered. If the SIC receiver uses a linear ZF equalizer, theneach transmitted symbol stream may be isolated by filtering the N_(R)modified symbol streams with a filter matched to that transmitted symbolstream. For simplicity, the following description assumes that thetransmitted symbol streams are recovered in an ascending order (i.e.,the symbol stream from transmit antenna 1 is recovered first, the symbolstream from transmit antenna 2 is recovered next, and so on, and symbolstream from transmit antenna N_(T) is recovered last). However, this isnot a requirement and the transmitted symbol streams may also berecovered in some other order.

The match filter for the l-th symbol stream to be recovered in the l-thstage has a unit norm vector, w _(l)(k), of N_(R) filter coefficientsfor each tone k, where k=0, 1, . . . (N_(F)−1). To minimize theinterference from the other (N_(T)−l) not-yet-recovered symbol streamson the l-th symbol stream, the vector w _(l)(k) is defined to beorthogonal to {h _(j)(k)} for j=l+1, l+2, . . . N_(T). This conditionmay be expressed as${{{{\underset{\_}{w}}_{l}^{H}(k)}{{\underset{\_}{h}}_{j}(k)}} = 0},$for j=l+1, l+2, . . . N_(T) and also for each k=0, 1, . . . (N_(F)−1).Since the transmitted symbol streams from the other (l−1) transmitantennas have already been recovered in prior stages and have beencanceled from the modified symbol streams y ^(l)(k) for the l-th stage,the vector w _(l)(k) does not need to be orthogonal to {h _(j)(k)}, forj=1, 2, . . . l−1 and k=0, 1, . . . (N_(F)−1).

The match filter response w _(l)(k) may be derived based on variousspatial or space-time processing techniques. For example, the matchfilter response w _(l)(k) may be derived using a linear ZF equalizer.For the SIC receiver, the channel response matrix, H(k), is reduced byone column in each stage as a transmitted symbol stream is recovered.For the l-th stage, the reduced channel response matrix, H ^(l)(k), isan (N_(R)×(N_(T)−l+1)) matrix, with (l−1) columns for the transmitantennas of the (l−1) prior-recovered symbol streams removed from theoriginal matrix H(k). The ZF equalizer response matrix${\underset{\_}{W}}_{ZF}^{l}(k)$for the l-th stage may be derived based on the reduced channel responsematrix, H ^(l)(k), as shown in equation (16). However, since H ^(l)(k)is different for each stage, ${\underset{\_}{W}}_{ZF}^{l}(k)$is also different for each stage. The match filter response w _(l)(k)for the l-th symbol stream recovered in the l-th stage may be expressedas ${{{\underset{\_}{w}}_{l}(k)} = {{\underset{\_}{w}}_{ZF}^{l}(k)}},$where ${\underset{\_}{w}}_{ZF}^{l}(k)$corresponds to l-th transmit antenna and is the first column of the ZFequalizer response matrix ${{\underset{\_}{W}}_{ZF}^{l}(k)},$which is derived for the l-th stage.

The detected symbol stream, {circumflex over (x)}_(l), for the l-thtransmit antenna may then be estimated as: $\begin{matrix}{{{\hat{x}}_{l}(k)} = {{{{\underset{\_}{w}}_{l}^{H}(k)}{{\underset{\_}{y}}^{l}(k)}} = {{{{\underset{\_}{w}}_{l}^{H}(k)}{{\underset{\_}{h}}_{l}(k)}{x_{l}(k)}} + {{{\underset{\_}{w}}_{l}^{H}(k)}{\underset{\_}{n}.}}}}} & {{Eq}\quad(18)}\end{matrix}$The spatial or space-time processing for the l-th stage of the SICreceiver can provide (N_(T)−l+1) detected symbol streams, {{circumflexover (x)}_(j)} for j=l, l+1, . . . N_(T). Each detected symbol streamincludes estimates of the modulation symbols transmitted on all N_(F)frequency subchannels of a respective transmit antenna. The spatialprocessing thus effectively maps the MIMO system to a number of parallelSISO systems. Of the (N_(T)−l+1) symbol streams detected at the l-thstage, the one corresponding to the l-th transmit antenna is selectedfor further processing to obtain the data for that symbol stream.

If the symbol streams can be recovered without error (or with minimalerrors) and if the channel response estimate is reasonably accurate,then cancellation of the interference due to the recovered symbolstreams is effective. The later recovered symbol streams would thenexperience less interference and may be able to achieve higher SNRs. Inthis way, higher performance may be achieved for all recovered symbolstreams (possibly except for the first recovered symbol stream). The SICprocessing technique can outperform the spatial/space-time receiverprocessing techniques if the interference due to each recovered streamcan be accurately estimated and canceled. This requires error-free orlow-error recovery of the transmitted symbol streams, which can beachieved in part by the use of an error-correction code for the symbolstream.

Typically, an important consideration for a SIC receiver is the order inwhich the transmitted symbol streams are detected. If the same data rateis used for all transmit antennas, then the detected symbol stream thatattains the highest SNR may be selected for recovery. However, with therate control described herein, the rates for the transmit antennas maybe selected such that all detected symbol streams are similarlyreliable. With rate control, the order in which the symbol streams aredetected is not an important consideration.

In an aspect, in a multi-channel system that employs multipletransmission channels for data transmission, each independentlyprocessed data stream may be modeled with an equivalent SISO system.Rate control may then be performed for each data stream in similarmanner as that described above for the SISO system.

MIMO-OFDM System with AAP

If AAP is used at the transmitter of a MIMO-OFDM system, then for eachtransmission symbol period the spatial or space-time processing at thereceiver provides N_(T) detected OFDM symbols that have been transmittedfrom the N_(T) transmit antennas. Each detected OFDM symbol comprisesN_(F) modulation symbols for the N_(F) frequency subchannels. The N_(T)detected OFDM symbols typically fade independently, and each OFDM symbolis distorted by the response of the spatial subchannel via which theOFDM symbol was received.

For the AAP scheme, interleaving is done in both the frequency and spacedomains. Hence, a codeword may be interleaved across all N_(T) detectedOFDM symbols. A MIMO-OFDM system with AAP (which employs all N_(T)N_(F)transmission channels to transmit a codeword) may then be mapped to anequivalent SISO system that employs N_(T)N_(F) subcarriers and occupiesN_(T) times the bandwidth of one spatial subchannel (and hence,encountering a channel of N_(T)L multipaths). If the mapping iseffective, then the equivalent SNR for the equivalent SISO system withan AWGN channel may then be used to select the proper data rate for theMIMO-OFDM system with a multipath channel.

FIG. 5A is a diagram illustrating the spectral efficiencies of thespatial subchannels in a MIMO-OFDM system with a multipath channel. Forthe MIMO-OFDM system, there are N_(T) spatial subchannels if the channelresponse matrix H(k) is full-rank (i.e., N_(S)=N_(T)≦N_(R)). In thiscase, each spatial subchannel is associated with a respective transmitantenna and has a bandwidth of W. The channel response of each spatialsubchannel (or each transmit antenna) is defined by h _(j)(k) for j=1,2, . . . N_(T) and k=0, 1, . . . (N_(F)−1), where h _(j)(k) is onecolumn of the matrix H(k) and includes N_(R) elements for the N_(R)receive antennas.

For each transmit antenna with channel response of h _(j)(k) and noisevariance of N₀, a plot 510 of the spectral efficiencies for the N_(F)frequency subchannels may be derived based on the constrained orunconstrained spectral efficiency function as shown in equation (2) or(3). An average spectral efficiency S_(avg) for each transmit antennamay also be derived as shown in equation (5). As shown in FIG. 5A, thespectral efficiency plots 510 a through 510 t for the N_(T) transmitantennas (or N_(T) spatial subchannels) may be different because ofindependent fading for these spatial subchannels.

FIG. 5B is a diagram illustrating the spectral efficiency of anequivalent SISO system used to model the MIMO-OFDM system shown in FIG.5A. The equivalent SISO system is defined to have an AWGN channel and aspectral efficiency equal to the average spectral efficiency of theMIMO-OFDM system being modeled. For the MIMO-OFDM system with N_(T)parallel colored-noise channels, each occupying a bandwidth of W, theoverall capacity C_(mimo) may be expressed as: $\begin{matrix}{{C_{mimo} = {W\quad{\log_{2}\left( \frac{{\Sigma_{s} + \Sigma_{n}}}{\Sigma_{n}} \right)}}},} & {{Eq}\quad(19)}\end{matrix}$where |Σ| is the determinant of Σ, and Σ_(s) is a diagonal matrix withthe post equalizer signal powers. The diagonal matrix Σ_(s) may bederived based on equation (18) and may be expressed as: $\begin{matrix}{\Sigma_{s} = {\begin{bmatrix}{{{\underset{\_}{w}}_{1}^{H}{\underset{\_}{h}}_{1}}}^{2} & 0 & \ldots & 0 \\0 & {{{\underset{\_}{w}}_{2}^{H}{\underset{\_}{h}}_{2}}}^{2} & \ldots & 0 \\\vdots & \vdots & ⋰ & \vdots \\0 & 0 & \ldots & {{{\underset{\_}{w}}_{N_{T}}^{H}{\underset{\_}{h}}_{N_{T}}}}^{2}\end{bmatrix}.}} & {{Eq}\quad(20)}\end{matrix}$The capacity C_(mimo) of the MIMO-OFDM system may then be expressed as:$\begin{matrix}{{C_{mimo} \geq {W\quad{\sum\limits_{j = 1}^{N_{T}}S_{j}}}},} & {{Eq}\quad(21)}\end{matrix}$where S_(j) is the spectral efficiency in bits/s/Hz corresponding to thej-th transmit antenna.For simplicity, the lower bound in equation (21), i.e.,${C_{mimo} = {W\quad{\sum\limits_{j = 1}^{N_{T}}S_{j}}}},$is used for the following description. However, the actual capacity ofthe MIMO-OFDM system may also be used, and this is within the scope ofthe invention.

The capacity C_(siso) of the equivalent SISO system occupying abandwidth of N_(T)W may be expressed as:C_(siso)=N_(T)WS_(equiv),  Eq (22)where S_(equiv) is the spectral efficiency in bits/s/Hz of theequivalent SISO system with AWGN channel.

Setting C_(siso) equal to C_(mimo) and combining equations (22) and(23), the spectral efficiency S_(equiv) of the equivalent SISO systemmay be expressed as: $\begin{matrix}{S_{equiv} = {\frac{\sum\limits_{j = 1}^{N_{T}}S_{j}}{N_{T}}.}} & {{Eq}\quad(23)}\end{matrix}$

The spectral efficiency S_(j) for each transmit antenna in the MIMO-OFDMsystem may be expressed as: $\begin{matrix}{{S_{j} = \frac{\sum\limits_{k = 0}^{N_{F} - 1}{f\left( {{{\underset{\_}{h}}_{j}(k)},{{\underset{\_}{w}}_{j}(k)},N_{0},{M(r)}} \right)}}{N_{F}}},} & {{Eq}\quad(24)}\end{matrix}$where w_(j)(k) is the ZF equalizer response for the j-th transmitantenna, e.g., the j-th column in the matrix W _(ZF)(k) determined inequation (16).

The function ƒ(x) in equation (24) is a function of SNR(k) andmodulation scheme M(r). The SNR for the k-th frequency subchannel of thej-th transmit antenna may be expressed as: $\begin{matrix}{{{SNR}_{j}(k)} = {\frac{\left| {{w_{j}^{H}(k)}{h_{j}(k)}} \right|^{2}}{N_{0}}.}} & {{Eq}\quad(25)}\end{matrix}$

The average spectral efficiency S_(avg, AAP) for the MIMO-OFDM systemwith AAP may then be expressed as: $\begin{matrix}{S_{{avg},{AAP}} = {\frac{\sum\limits_{j = 1}^{N_{T}}{\sum\limits_{k = 0}^{N_{F} - 1}{f\left( {{{\underset{\_}{h}}_{j}(k)},{{\underset{\_}{w}}_{j}(k)},N_{0},{M(r)}} \right)}}}{N_{T}N_{F}}.}} & {{Eq}\quad(26)}\end{matrix}$The average spectral efficiency S_(avg, AAP) for the MIMO-OFDM systemwith AAP is then used as the spectral efficiency S_(equiv) of theequivalent SISO system (i.e., S_(equiv)=S_(avg, AAP))

The equivalent SNR for the spectral efficiency S_(equiv) in theequivalent SISO system may then be determined for the MIMO-OFDM systemwith AAP, as shown in equation (6), which is:Ψ=SNR _(equiv) =g(S _(equiv) , M(r)),  Eq (27)As shown in equation (27), the equivalent SNR is obtained for theequivalent system spectral efficiency S_(equiv), which as shown inequations (24) and (26) is obtained by averaging the spectralefficiencies S_(j), for j=1, 2, . . . N_(T), of all N_(T) transmitantennas. The spectral efficiency S_(j) of each transmit antenna is inturn obtained by averaging the spectral efficiencies S_(j)(k) of allN_(F) frequency subchannels. The equivalent SNR is thus determined bythe average spectral efficiency of all frequency subchannels and spatialsubchannels, as shown in FIG. 5B. The equivalent SNR may then be used asthe metric Ψ to determine the rate for the data transmission over alltransmit antennas, in similar manner as that described above for theSISO system.

As shown in FIG. 5B, a discontinuity may exist in the spectralefficiency distribution plot 520 for the equivalent SISO system due topiecewise concatenation of the spectral efficiency functions ƒ_(j)(x),for j=1, 2, . . . N_(T), for the N_(T) transmit antennas. However, thisdiscontinuity effect is mitigated by the role of the interleaver used atthe transmitter to interleave data prior to transmission across thefrequency and space domains.

MIMO-OFDM System with PAP

If PAP is used at the transmitter of a MIMO-OFDM system, then the ratecontrol may be performed for each of the N_(T) data streams transmittedfrom the N_(T) transmit antennas. At the receiver, eitherspatial/space-time processing or SIC processing may be used to recoverthe N_(T) transmitted symbol streams. Since the SIC processing mayprovide improved performance over the spatial/space-time processing forPAP, the following description is for a SIC receiver.

For the SIC receiver, to recover the symbol stream from the l-thtransmit antenna in the l-th stage, the interference from the (l−1)prior-recovered symbol streams are assumed to be canceled, and theinterference from the other (N_(T)−l) not-yet-recovered symbol streamsmay be minimized (or nulled out) by selecting the proper match filterresponse w _(l)(k) for the symbol stream to be recovered in this stage.The match filter response w _(l)(k) includes N_(R) elements for theN_(R) receive antennas, with each element being a vector of N_(F)coefficients for the N_(F) frequency subchannels. Thus, each stage ofthe SIC receiver resembles a (1, N_(R)) SIMO system.

The average spectral efficiency S_(avg, PAP,l) for each transmit antennain the MIMO-OFDM system with PAP may be expressed as: $\begin{matrix}{{S_{{avg},{PAP},l} = \frac{\sum\limits_{k = 0}^{N_{F} - 1}{f\left( {{{\underset{\_}{h}}_{l}(k)},{{\underset{\_}{w}}_{l}(k)},N_{0},{M(r)}} \right)}}{N_{F}}},} & {{Eq}\quad(28)}\end{matrix}$where h _(l)(k) and w _(l)(k) are respectively the channel response andthe filter response associated with the l-th transmit antenna. Theaverage spectral efficiency S_(avg, PAP,l) for each transmit antenna inthe MIMO-OFDM system with PAP is used as the spectral efficiencyS_(equiv) of the equivalent SISO system (i.e., S_(equiv)=S_(avg, PAP,l))to determine the rate for the transmit antenna.

The function ƒ(x) in equation (28) is a function of SNR and modulationscheme M(r). The SNR for the k-th frequency subchannel of the l-thtransmit antenna may be expressed as: $\begin{matrix}{{{SNR}_{l}(k)} = {\frac{\left| {{{\underset{\_}{w}}_{l}^{H}(k)}{{\underset{\_}{h}}_{l}(k)}} \right|^{2}}{N_{0}}.}} & {{Eq}\quad(29)}\end{matrix}$As noted above, the match filter response w _(l)(k) for the symbolstream recovered in the l-th stage is a column of the ZF equalizerresponse matrix ${{\underset{\_}{W}}_{ZF}^{l}(k)}.$The matrix ${\underset{\_}{W}}_{ZF}^{l}(k)$is derived for the l-th stage based on the reduced channel responsematrix, H ^(l)(k), which has (l−1) columns for the (l−1) prior-recoveredsymbol streams removed.

For each transmit antenna in the MIMO-OFDM system with PAP, the spectralefficiency S_(equiv) of the equivalent SISO system may be determined asshown in equation (28), and the equivalent SNR may then be determinedfor the spectral efficiency S_(equiv) as shown in equation (27). Theequivalent SNR for each transmit antenna is determined by the averagespectral efficiency of all frequency subchannels of the transmitantenna, as shown in FIG. 5A. The equivalent SNR for each transmitantenna may then be used as the metric Ψ to determine the rate for thattransmit antenna, in similar manner as that described above for the SISOsystem.

Multi-channel System with MCP

For a multi-channel processing (MCP) scheme, one or more data streamsare independently processed (e.g., encoded, interleaved, and modulated)at the transmitter to provide one or more corresponding symbol streams,and each symbol stream may then be transmitted over a respective groupof transmission channels. Each transmission channel group may include(1) some or all frequency subchannels of a spatial subchannel, (2) someor all frequency subchannels of multiple spatial subchannels, (3) someor all spatial subchannels of a frequency subchannel, (4) some or allspatial subchannels of multiple frequency subchannels, (5) anycombination of transmission channels, or (6) all transmission channels.The rate for each independently processed data stream may be controlledsuch that improved performance (e.g., high throughput) is achieved. TheAAP and PAP may be viewed as variants of the MCP scheme.

FIG. 6 is a flow diagram of an embodiment of a process 600 forcontrolling the rate of one or more independently processed datastreams, each of which is transmitted over a respective group oftransmission channels.

Initially, the first data stream to be rate controlled is selected, forexample, by setting a variable m used to denote the data stream to one(i.e., m=1) (step 612). The group of transmission channels used for datastream d_(m) is then determined (step 614). For the AAP scheme, one datastream is transmitted over all frequency subchannels of all spatialsubchannels, and the transmission channel group would then include alltransmission channels. For the PAP scheme, one data stream istransmitted over all frequency subchannels of each spatial subchannel,and the transmission channel group would then include all frequencysubchannels for the transmit antenna used for data stream d_(m). For theMCP scheme, the transmission channel group may include any combinationof frequency and spatial subchannels.

The highest available rate R_(m)(r) that may be used for data streamd_(m) is then selected for evaluation (step 616). If the available ratesare included in a set in increasing order, then the highest availablerate may be selected by setting a variable r to P (i.e., r=P), which isthe highest index for the set. The same rate set may be used for alldata streams, or each data stream may be associated with a differentrate set.

Parameters associated with data stream d_(m) and rate R_(m)(r) are thendetermined (step 618). Some parameters may relate to the processing fordata stream d_(m), such as the modulation scheme M_(m)(r) to be used forthe data stream. Some other parameters may relate to the communicationchannel, such as the channel response h_(i,j)(k) for each transmissionchannel in the group and the noise variance N₀.

A metric Ψ is then determined for data stream d_(m) (block 620). In anembodiment, the metric Ψ relates to the SNR for an equivalent SISOsystem that models the group of transmission channels used for datastream d_(m). The metric Ψ may be obtained by first determining theaverage spectral efficiency S_(avg, MCP,m) of all transmission channelsused for data stream d_(m) (step 622), which may be expressed as:$\begin{matrix}{{S_{{avg},{MCP},m} = \frac{\sum\limits_{n = 0}^{N_{m}}{f\left( {h_{n},w_{n},N_{0},{M_{m}(r)}} \right)}}{N_{m}}},} & {{Eq}\quad(30)}\end{matrix}$where h_(n) and w_(n) are respectively the channel response and filterresponse associated with the n-th transmission channel, where n is anindex comprising (i,j,k), M_(m)(r) is the modulation scheme used fordata stream d_(m), and N_(m) is the number of transmission channels usedfor data stream d_(m). For data stream d_(m), the same modulation schememay be used for all transmission channels, as shown in equation (30), ordifferent modulation schemes may be used for different transmissionchannels.

The spectral efficiency of the equivalent SISO system is then set equalto the average spectral efficiency of the transmission channels used fordata stream d_(m) (i.e., S_(equiv,m)=S_(avg, MCP,m)) (step 624). Theequivalent SNR needed to support a rate of S_(equiv,m) in the equivalentSISO system is then determined based on equation (27) (step 626). Theequivalent SNR may be adjusted by a back-off amount to account forimplementation losses, as described above for the SISO system (step628). This step is optional and represented by a dashed box for step628. The metric Ψ is then set equal to the unadjusted or adjustedequivalent SNR (step 630). The SNR required to reliably transmit datastream d_(m) at rate R_(m)(r) for the multi-channel system with an AWGNchannel is then determined, e.g., from a table (step 632).

A determination is then made whether or not rate R_(m)(r) is supportedby the group of transmission channels used for data stream d_(m)(step636). If the metric Ψ is greater than or equal to the required SNR(i.e., Ψ≧SNR_(req)), then rate R_(m)(r) is deemed to be supported fordata stream d_(m), and the process proceeds to step 640. Otherwise, thenext lower available rate is selected for data stream d_(m) bydecrementing the index r (i.e., r=r−1) (step 638). The process thenreturns to step 618 to evaluate the new rate.

At step 640, a determination is made whether or not rate control hasbeen performed for all data streams. If the answer is no, then ratecontrol is performed for the next data stream by incrementing thevariable m (i.e., m=m+1) (step 642). The process then returns to step614 to determine the rate for the new data stream d_(m). Otherwise, ifrate control has been performed for all data streams, then the set ofrates {R_(m)(r)}, for m=1, 2, . . . , N_(D), to be used for the N_(D)independently processed data streams is provided (step 644). The processthen terminates.

It can be shown via computer simulation that the rate control techniquesdescribed herein can approach the performance of an optimal rateselection scheme. The optimal selection scheme is a non-practical schemethat tests every available rate (for a given channel realization) andselects the highest rate whose PER conforms to the desired PER of P_(e).The rate control techniques described herein may thus be used toimplement a realizable rate control scheme having high performance.

FIG. 7 is a block diagram of an embodiment of a transmitter system 110 aand a receiver system 150 a in multi-channel communication system 100.

At transmitter system 110 a, traffic data is provided from a data source708 to a TX data processor 710. TX data processor 710 may demultiplexthe data into a number of data streams, and further formats, codes, andinterleaves each data stream based on a coding scheme to provide acorresponding coded data stream. The data rate and the coding for eachdata stream may be determined by a data rate control and a codingcontrol, respectively, provided by a controller 730.

The coded data is then provided to a modulator 720, which may alsoreceive pilot data (e.g., data used for channel estimation and otherfunctions). The pilot data may be multiplexed with the coded trafficdata, e.g., using time division multiplex (TDM) or code divisionmultiplex (CDM), in all or a subset of the transmission channels used totransmit the traffic data. For OFDM, the processing by modulator 720 mayinclude (1) modulating the received data with one or more modulationschemes, (2) transforming the modulated data to form OFDM symbols, and(3) appending a cyclic prefix to each OFDM symbol to form acorresponding transmission symbol. The modulation is performed based ona modulation control provided by controller 730. A transmission symbolstream is then provided to each transmitter (TMTR) 722.

Each transmitter 722 converts the received transmission symbol streaminto one or more analog signals and further conditions (e.g., amplifies,filters, and upconverts) the analog signals to generate a modulatedsignal suitable for transmission over the communication channel. Themodulated signal from each transmitter 722 is then transmitted via anassociated antenna 724 to the receiver system.

At receiver system 150 a, the transmitted modulated signals are receivedby each of antennas 752 a through 752 r, and the received signal fromeach antenna is provided to an associated receiver (RCVR) 754. Eachreceiver 754 conditions (e.g., filters, amplifies, and downconverts) itsreceived signal and digitizes the conditioned signal to provide datasamples. The sample streams from receivers 754 a through 754 r are thenprovided to a receiver processor 760, which includes a demodulator 762and an RX data processor 764.

For OFDM, the processing by demodulator 762 may include (1) removing thecyclic prefix previously appended to each OFDM symbol, (2) transformingeach recovered OFDM symbol, and (3) demodulating the recoveredmodulation symbols in accordance with one or more demodulation schemescomplementary to the one or more modulation schemes used at thetransmitter system. RX data processor 764 then decodes the demodulateddata to recover the transmitted traffic data. The processing bydemodulator 762 and RX data processor 764 is complementary to thatperformed by modulator 720 and TX data processor 710, respectively, attransmitter system 110 a.

As shown in FIG. 7, demodulator 762 may derive estimates of the channelcharacteristics (e.g., the channel response and noise variance) andprovide these channel estimates to a controller 770. RX data processor764 may also derive and provide the status of each received packet andmay further provide one or more other performance metrics indicative ofthe decoded results. Based on the various types of information receivedfrom demodulator 762 and RX data processor 764, controller 770 maydetermine or select a particular rate for each independently processeddata stream based on the techniques described above. Feedbackinformation in the form of a set of selected rates for the data streams,the channel response estimates, ACK/NACK for the receive packet, and soon, or any combination thereof, may be provided by controller 770,processed by a TX data processor 778, modulated by a modulator 780, andconditioned by transmitters 754, and transmitted by antennas 752 back totransmitter system 110 a.

At transmitter system 110 a, the modulated signals from receiver system150 a are received by antennas 724, conditioned by receivers 722,demodulated by a demodulator 740, and processed by a RX data processor742 to recover the feedback information transmitted by the receiversystem. The feedback information is then provided to controller 730 andused to control the processing of the data streams. For example, thedata rate of each data stream may be determined based on the selectedrate provided by the receiver system, or may be determined based on thechannel estimates from the receiver system. The specific coding andmodulation schemes associated with the selected rate are determined andreflected in the coding and modulation controls provided to TX dataprocessor 710 and modulator 720. The received ACK/NACK may be used toinitiate an incremental transmission whereby a small portion of a packetreceived in error is retransmitted to allow the receiver to correctlyrecover the packet.

Controllers 730 and 770 direct the operation at the transmitter andreceiver systems, respectively. Memories 732 and 772 provide storage forprogram codes and data used by controllers 730 and 770, respectively.

FIG. 8 is a block diagram of a transmitter unit 800, which is anembodiment of the transmitter portion of transmitter system 110 a inFIG. 7. Transmitter unit 800 includes (1) a TX data processor 710 a thatcodes each data stream in accordance with a particular coding scheme toprovide a corresponding coded data stream and (2) a modulator 720 a thatmodulates and performs OFDM processing on the coded data streams toprovide transmission symbol streams.

In an embodiment, each data stream may be associated with it own datarate and coding and modulation schemes, which are identified by thecontrols provided by controller 730. The rate selection for each datastream may be performed as described above.

In the embodiment shown in FIG. 8, TX data processor 710 a includes ademultiplexer 810, N_(D) encoders 812 a through 812 s, and N_(D) channelinterleavers 814 a through 814 s (i.e., one set of encoder and channelinterleaver for each data stream). Demultiplexer 810 demultiplexes thetraffic data (i.e., the information bits) into N_(D) data streams, whereN_(D) an be any integer one or greater. The N_(D) data streams areprovided at data rates determined to be supported by the N_(D) groups oftransmission channels used for these data streams. Each data stream isprovided to a respective encoder 812.

Each encoder 812 codes a respective data stream based on the specificcoding scheme selected for that data stream to provide coded bits. Thecoding increases the reliability of the data transmission. The codingscheme may include any combination of cyclic redundancy check (CRC)coding, convolutional coding, Turbo coding, block coding, and so on. Thecoded bits from each encoder 812 are then provided to a respectivechannel interleaver 814, which interleaves the coded bits based on aparticular interleaving scheme. The interleaving provides time diversityfor the coded bits, permits the data to be transmitted based on anaverage SNR for the transmission channels used for the data stream,combats fading, and further removes correlation between coded bits usedto form each modulation symbol. The N_(D) coded data streams are thenprovided to modulator 720 a.

In the embodiment shown in FIG. 8, modulator 720 a includes N_(D) symbolmapping elements 822 a through 822 s (one for each data stream), amultiplexer/demultiplexer 824, and N_(T) OFDM modulators (one for eachtransmit antenna), with each OFDM modulator including an inverse Fouriertransform (IFFT) unit 826 and a cyclic prefix generator 828.

Each symbol mapping element 822 receives a respective coded data streamand maps the coded and interleaved bits based on the modulation schemeselected for that data stream to form modulation symbols. Each symbolmapping element 822 groups each set of q_(m) coded and interleaved bitsto form a non-binary symbol, and further maps the non-binary symbol to aspecific point in a signal constellation corresponding to the selectedmodulation scheme (e.g., QPSK, M-PSK, or M-QAM). Each mapped signalpoint corresponds to an M_(m)-ary modulation symbol, where M_(m)corresponds to the specific modulation scheme selected for data streamd_(m) and M_(m)=2^(q) ^(m) . Pilot data may also be symbol mapped toprovide pilot symbols, which may then be multiplexed (e.g., using TDM orCDM) with the modulation symbols for the traffic data. Symbol mappingelements 822 a through 822 s then provide the modulation symbols for theN_(D) data streams to multiplexer/demultiplexer 824.

Each data stream is transmitted on a respective group of transmissionchannels, and each transmission channel group may include any number andcombination of spatial and frequency subchannels.Multiplexer/demultiplexer 824 provides the modulation symbols for eachdata stream to the transmission channels to be used for that datastream. Multiplexer/demultiplexer 824 then provides N_(T) modulationsymbol streams to the N_(T) OFDM modulators.

For the AAP scheme, one data stream is transmitted over all transmissionchannels, and only one set of encoder 812, channel interleaver 814, andsymbol mapping element 822 is needed. Multiplexer/demultiplexer 824 thendemultiplexes the modulation symbols into N_(T) modulation symbolstreams for the N_(T) transmit antennas.

For the PAP scheme, one data stream is transmitted over all frequencysubchannels of each transmit antenna, and N_(T) sets of encoder 812,channel interleaver 814, and symbol mapping element 822 are provided(i.e., N_(D)=N_(S)). Multiplexer/demultiplexer 824 then simply passesthe modulation symbols from each symbol mapping element 822 to anassociated IFFT 826.

For the MCP scheme, each data stream is transmitted over a respectivegroup of transmission channels. Multiplexer/demultiplexer 824 performsthe appropriate multiplexing/demultiplexing of the modulation symbols tothe proper transmission channels.

Within each OFDM modulator, IFFT 826 receives the modulation symbolstream, groups each set of N_(F) modulation symbols to form acorresponding modulation symbol vector, and converts this vector intoits time-domain representation (which is referred to as an OFDM symbol)using the inverse fast Fourier transform. For each OFDM symbol, cyclicprefix generator 828 repeats a portion of the OFDM symbol to form acorresponding transmission symbol. The cyclic prefix ensures that thetransmission symbol retains its orthogonal properties in the presence ofmultipath delay spread, thereby improving performance againstdeleterious path effects such as channel dispersion caused by frequencyselective fading. Cyclic prefix generator 828 then provides a stream oftransmission symbols to an associated transmitter 722.

Each transmitter 722 receives and processes a respective transmissionsymbol stream to generate a modulated signal, which is then transmittedfrom the associated antenna 724.

The coding and modulation for MIMO systems with and without OFDM aredescribed in further detail in the following U.S. patent applications:

-   -   U.S. patent application Ser. No. 09/993,087, entitled        “Multiple-Access Multiple-Input Multiple-Output (MIMO)        Communication System,” filed Nov. 6, 2001;    -   U.S. patent application Ser. No. 09/854,235, entitled “Method        and Apparatus for Processing Data in a Multiple-Input        Multiple-Output (MIMO) Communication System Utilizing Channel        State Information,” filed May 11, 2001;    -   U.S. patent application Ser. Nos. 09/826,481 and 09/956,449,        both entitled “Method and Apparatus for Utilizing Channel State        Information in a Wireless Communication System,” respectively        filed Mar. 23, 2001 and Sep. 18, 2001;    -   U.S. patent application Ser. No. 09/776,075, entitled “Coding        Scheme for a Wireless Communication System,” filed Feb. 1, 2001;        and    -   U.S. patent application Ser. No. 09/532,492, entitled “High        Efficiency, High Performance Communications System Employing        Multi-Carrier Modulation,” filed Mar. 30, 2000.        These applications are all assigned to the assignee of the        present application and incorporated herein by reference. Other        designs for the transmitter unit may also be implemented and are        within the scope of the invention.

FIG. 9 is a block diagram of an embodiment of a receiver processor 760a, which is one embodiment of receiver processor 760 in FIG. 7. Thetransmitted modulated signals are received by antennas 752 and processedby receivers 754 to provide N_(R) sample streams, which are thenprovided to an RX OFDM processor 910 within demodulator 762 a.

Within demodulator 762 a, each sample stream is provided to a respectiveOFDM demodulator, which includes a cyclic prefix removal element 912 andan FFT unit 914. Element 912 removes the cyclic prefix included in eachtransmission symbol to provide a corresponding recovered OFDM symbol.FFT 914 then transforms each recovered OFDM symbol using the fastFourier transform to provide a vector of N_(F) recovered modulationsymbols for the N_(F) frequency subchannels for each transmission symbolperiod. FFT units 914 a through 914 r provide N_(R) received symbolstreams to a spatial processor 920.

Spatial processor 920 performs spatial or space-time processing on theN_(R) received symbol streams to provide N_(T) detected symbol streams,which are estimates of the N_(T) transmitted symbol streams. Spatialprocessor 920 may implement a linear ZF equalizer, a channel correlationmatrix inversion (CCMI) equalizer, a minimum mean square error (MMSE)equalizer, an MMSE linear equalizer (MMSE-LE), a decision feedbackequalizer (DFE), or some other equalizer, which are described in detailin the aforementioned U.S. patent application Ser. Nos. 09/993,087,09/854,235, 09/826,481, and 09/956,44.

A multiplexer/demultiplexer 922 then multiplexes/demultiplexes thedetected symbols, and provides N_(D) aggregated detected symbol streamsfor the N_(D) data streams to N_(D) symbol demapping elements 924. Eachsymbol demapping element 924 then demodulates the detected symbols inaccordance with a demodulation scheme that is complementary to themodulation scheme used for the data stream. The N_(D) demodulated datastreams from N_(D) symbol demapping elements 924 are then provided to aRX data processor 764 a.

Within RX data processor 764 a, each demodulated data stream isde-interleaved by a channel de-interleaver 932 in a manner complementaryto that performed at the transmitter system for the data stream, and thede-interleaved data is further decoded by a decoder 934 in a mannercomplementary to that performed at the transmitter system. For example,a Turbo decoder or a Viterbi decoder may be used for decoder 934 ifTurbo or convolutional coding, respectively, is performed at thetransmitter unit. The decoded data stream from each decoder 934represents an estimate of the transmitted data stream. Decoder 934 mayalso provide the status of each received packet (e.g., indicatingwhether it was received correctly or in error). Decoder 934 may furtherstore demodulated data for packets not decoded correctly, so that thisdata may be combined with data from a subsequent incrementaltransmission and decoded.

In the embodiment shown in FIG. 9, a channel estimator 940 estimates thechannel response and the noise variance and provides these estimates tocontroller 770. The channel response and noise variance may be estimatedbased on the detected symbols for the pilot.

Controller 770 may be designed to perform various functions related torate selection. For example, controller 770 may determine the maximumdata rate that may be used for each data stream based on the channelestimates and other parameters such as the modulation scheme.

FIG. 10 is a block diagram of an embodiment of a receiver processor 760b, which is another embodiment of receiver processor 760 in FIG. 7.Receiver processor 760 b performs SIC processing and may be used if thePAP or MCP scheme is employed at the transmitter system. For simplicity,the following description for receiver processor 760 b assumes that thePAP scheme is employed.

In the embodiment shown in FIG. 10, receiver processor 760 b includes(1) RX OFDM processor 910 that processes the N_(R) sample streams toprovide N_(R) received symbol streams, as described above, and (2) aspatial/data processor 1000. Spatial/data processor 1000 includes anumber of successive (i.e., cascaded) receiver processing stages 1010 athrough 1010 t, one stage for each of the symbol streams to berecovered. Each receiver processing stage 1010 (except for the laststage 1010 t) includes a spatial processor 1020, an RX data processor1030, and an interference canceller 1040. The last stage 1010 t includesonly spatial processor 1020 t and RX data processor 1030 t.

For the first stage 1010 a, spatial processor 1020 a receives andprocesses the N_(R) received symbol streams (denoted as a vector y ¹)from RX OFDM processor 910 based on a particular spatial or space-timeequalizer (e.g., a linear ZF equalizer, a CCMI equalizer, an MMSEequalizer, a MMSE-LE, or a DFE) to provide N_(T) detected symbol streams(denoted as a vector

¹). One data stream is selected for recovery, and spatial processor 1020a provides the detected symbol stream {circumflex over (x)}₁ for thisdata stream to RX data processor 1030 a. Processor 1030 a furtherprocesses (e.g., demodulates, deinterleaves, and decodes) the selecteddetected symbol stream {circumflex over (x)}₁ to provide a correspondingdecoded data stream. Spatial processor 1020 a may further provide anestimate of the channel response, which is used to perform the spatialor space-time processing for all stages.

For the first stage 1010 a, interference canceller 1040 a receives theN_(R) received symbol streams from receivers 154 (i.e., the vector y ¹).Interference canceller 1040 a also receives and processes (e.g.,encodes, interleaves, and symbol maps) the decoded data stream from RXdata processor 1030 a to provide a remodulated symbol stream, {hacekover (x)}₁, which is an estimate of the symbol stream just recovered.The remodulated symbol stream {hacek over (x)}₁ is further processed inthe time or frequency domain to derive estimates of the interferencecomponents (denoted as an interference vector i ¹) due to thejust-recovered symbol stream. For the time-domain implementation, theremodulated symbol stream {hacek over (x)}₁ is OFDM processed to obtaina transmission symbol stream, which is further convolved by each ofN_(R) elements in a channel impulse response vector h ₁ to derive N_(R)interference components due to the just-recovered symbol stream. Thevector h ₁ is a column of the channel impulse response matrix, H,corresponding to transmit antenna 1 used for the just-recovered symbolstream. The vector h ₁ includes N_(R) elements that define the channelimpulse response between transmit antenna 1 and the N_(R) receiveantennas. For the frequency-domain implementation, the remodulatedsymbol stream {hacek over (x)}₁ is multiplied by each of N_(R) elementsin a channel frequency response vector h ₁ (which is a column of thematrix H) to derive N_(R) interference components. The interferencecomponents i ¹ are then subtracted from the first stage's input symbolstreams y ¹ to derive N_(R) modified symbol streams (denoted as a vectory ²), which include all but the subtracted (i.e., cancelled)interference components. The N_(R) modified symbol streams are thenprovided to the next stage.

For each of the second through last stages 1010 b through 1010 t, thespatial processor for that stage receives and processes the N_(R)modified symbol streams from the interference canceller in the precedingstage to derive the detected symbol streams for that stage. For eachstage, one detected symbol stream is selected and processed by the RXdata processor to provide the corresponding decoded data stream. Foreach of the second through second-to-last stages, the interferencecanceller in that stage receives the N_(R) modified symbol streams fromthe interference canceller in the preceding stage and the decoded datastream from the RX data processor within the same stage, derives theN_(R) interference components due to the symbol stream recovered by thatstage, and provides N_(R) modified symbol streams for the next stage.

The successive interference cancellation receiver processing techniqueis described in further detail in the aforementioned U.S. patentapplication Ser. Nos. 09/993,087 and 09/854,235.

FIGS. 7 and 9 show a simple design whereby the receiver sends back therates for the data streams. Other designs may also be implemented andare within the scope of the invention. For example, the channelestimates may be sent to the transmitter (instead of the rates), whichmay then determine the rates for the data streams based on these channelestimates.

The rate control techniques described herein may be implemented usingvarious designs. For example, channel estimator 940 in FIG. 9 used toderive and provide the channel estimates may be implemented by variouselements in the receiver system. Some or all of the processing todetermine the rate may be performed by controller 770 (e.g., with one ormore look-up tables stored in memory 772). Other designs for performingthe rate control may also be contemplated and are within the scope ofthe invention.

The rate control techniques described herein may be implemented byvarious means. For example, these techniques may be implemented inhardware, software, or a combination thereof. For a hardwareimplementation, some of the elements used to implement the rate controlmay be implemented within one or more application specific integratedcircuits (ASICs), digital signal processors (DSPs), digital signalprocessing devices (DSPDs), programmable logic devices (PLDs), fieldprogrammable gate arrays (FPGAs), processors, controllers,micro-controllers, microprocessors, other electronic units designed toperform the functions described herein, or a combination thereof.

For a software implementation, some portions of the rate control may beimplemented with modules (e.g., procedures, functions, and so on) thatperform the functions described herein. The software codes may be storedin a memory unit (e.g., memory 732 or 772 in FIG. 7) and executed by aprocessor (e.g., controller 730 or 770). The memory unit may beimplemented within the processor or external to the processor, in whichcase it can be communicatively coupled to the processor via variousmeans as is known in the art.

The previous description of the disclosed embodiments is provided toenable any person skilled in the art to make or use the presentinvention. Various modifications to these embodiments will be readilyapparent to those skilled in the art, and the generic principles definedherein may be applied to other embodiments without departing from thespirit or scope of the invention. Thus, the present invention is notintended to be limited to the embodiments shown herein but is to beaccorded the widest scope consistent with the principles and novelfeatures disclosed herein.

1. In a multi-channel communication system, a method for determining a rate for a data transmission over a wireless communication channel, comprising: identifying a plurality of transmission channels to be used for the data transmission; defining an equivalent system for the transmission channels based on one or more estimated channel characteristics of the transmission channels; deriving a metric for the transmission channels based on the equivalent system; and determining a particular rate for the data transmission based on the metric.
 2. The method of claim 1, further comprising: determining an average spectral efficiency of the transmission channels based on the one or more estimated channel characteristics, and wherein the equivalent system is defined to have an additive white Gaussian noise (AWGN) channel and a spectral efficiency equal to the average spectral efficiency of the transmission channels.
 3. The method of claim 2, further comprising: estimating a spectral efficiency of each transmission channel based on the one or more estimated channel characteristics, and wherein the average spectral efficiency of the transmission channels is determined based on the estimated spectral efficiencies of the transmission channels.
 4. The method of claim 3, wherein the spectral efficiency of each transmission channel is estimated based on a constrained spectral efficiency function.
 5. The method of claim 4, wherein the spectral efficiency of each transmission channel is further estimated based on a modulation scheme to be used for the data transmission.
 6. The method of claim 3, wherein the spectral efficiency of each transmission channel is estimated based on an unconstrained spectral efficiency function.
 7. The method of claim 2, wherein the deriving the metric includes determining an equivalent signal-to-noise-and-interference ratio (SNR) for the equivalent system, and wherein the metric is related to the equivalent SNR.
 8. The method of claim 7, wherein the equivalent SNR is determined based on an inverse function of a spectral efficiency function used to estimate a spectral efficiency of each transmission channel.
 9. The method of claim 7, wherein the deriving the metric further includes adjusting the equivalent SNR to account for losses in the communication system, and wherein the metric is related to the adjusted equivalent SNR.
 10. The method of claim 1, further comprising: determining a particular modulation scheme to use for the data transmission, and wherein the equivalent system is further defined based on the modulation scheme.
 11. The method of claim 1; further comprising: determining an SNR required to support the particular data rate by the communication system, and wherein the particular data rate is determined to be supported by the transmission channels if the required SNR is less than or equal to the metric.
 12. The method of claim 1, wherein the one or more estimated channel characteristics comprise an SNR for each transmission channel.
 13. The method of claim 1, wherein the one or more estimated channel characteristics comprise an estimated frequency response and an estimated noise variance for the transmission channels.
 14. The method of claim 1, wherein the transmission channels are frequency subchannels or spatial subchannels, or both, in a multipath wireless communication channel with frequency selective fading.
 15. The method of claim 1, wherein the multi-channel communication system is a multiple-input multiple-output (MIMO) communication system and the transmission channels correspond to spatial subchannels.
 16. The method of claim 1, wherein the multi-channel communication system is an orthogonal frequency division multiplex (OFDM) communication system and the transmission channels correspond to frequency subchannels.
 17. The method of claim 1, wherein the multi-channel communication system is a multiple-input multiple-output (MIMO) communication system that employs orthogonal frequency division multiplex (OFDM), and the transmission channels correspond to frequency subchannels of spatial subchannels.
 18. The method of claim 1, wherein a set of rates is available for the data transmission, the method further comprising: evaluating each of one or more available rates to determine a highest rate supported by the transmission channels.
 19. In a multi-channel communication system, a method for determining a rate for a data transmission over a wireless communication channel, comprising: identifying a group of transmission channels to be used for the data transmission; obtaining an estimated signal-to-noise-and-interference ratio (SNR) of each transmission channel; estimating spectral efficiency of each transmission channel based on the estimated SNR for the transmission channel; determining an average spectral efficiency of the transmission channels based on estimated spectral efficiencies of the transmission channels; determining an equivalent SNR for an equivalent system with a spectral efficiency equal to the average spectral efficiency of the transmission channels; determining a required SNR to support a particular data rate by the communication system; and determining whether the particular rate is supported by the transmission channels for the data transmission based on the equivalent SNR and the required SNR.
 20. The method of claim 19, wherein the spectral efficiency of each transmission channel is estimated based on an unconstrained spectral efficiency function.
 21. The method of claim 19, wherein the spectral efficiency of each transmission channel is further estimated based on a modulation scheme to be used for the data transmission.
 22. The method of claim 19, wherein the multi-channel communication system is a MIMO communication system that employs OFDM.
 23. In a multi-channel communication system, a method for determining a set of rates for a set of data streams to be transmitted over a wireless communication channel, comprising: identifying a group of transmission channels to be used for each data stream; defining an equivalent system for each transmission channel group based on one or more estimated channel characteristics of the transmission channels in the group; deriving a metric for each transmission channel group based on the associated equivalent system; and determining a rate for each data stream based on the metric associated with the data stream.
 24. The method of claim 23, further comprising: estimating a spectral efficiency of each transmission channel based on the one or more estimated channel characteristics, and determining an average spectral efficiency of the transmission channels in each group based on estimated spectral efficiencies of the transmission channels, and wherein the equivalent system for each transmission channel group is defined to have an additive white Gaussian noise (AWGN) channel and a spectral efficiency equal to the average spectral efficiency of the transmission channels in the group.
 25. The method of claim 24, wherein the spectral efficiency of each transmission channel is estimated based on an unconstrained or constrained spectral efficiency function.
 26. The method of claim 23, further comprising: for each data stream, determining an SNR required to support a particular rate by the communication system, and wherein the particular rate is determined to be supported by the group of transmission channels for the data stream if the required SNR is less than or equal to the metric associated with the data stream.
 27. The method of claim 23, wherein the multi-channel communication system is a MIMO communication system that employs OFDM, and the transmission channels correspond to frequency subchannels of spatial subchannels.
 28. The method of claim 27, wherein each data stream is transmitted over a respective transmit antenna, and each transmission channel group includes all frequency subchannels for one transmit antenna.
 29. A memory communicatively coupled to a digital signal processing device (DSPD) capable of interpreting digital information to: identify a plurality of transmission channels to be used for the data transmission; define an equivalent system for the transmission channels based on one or more estimated channel characteristics of the transmission channels; derive a metric for the transmission channels based on the equivalent system; and determine a particular rate for the data transmission based on the metric.
 30. The memory of claim 29, wherein the DSPD is further capable of interpreting digital information to: estimate a spectral efficiency of each transmission channel based on the one or more estimated channel characteristics, and determine an average spectral efficiency of the transmission channels based on estimated spectral efficiencies of the transmission channels, and wherein the equivalent system is defined to have an additive white Gaussian noise (AWGN) channel and a spectral efficiency equal to the average spectral efficiency of the transmission channels.
 31. A receiver unit in a multi-channel communication system, comprising: a channel estimator operative to derive estimates of one or more characteristics of a plurality of transmission channels; and a rate selector operative to define an equivalent system based on the one or more estimated channel characteristics of the transmission channels, derive a metric for the transmission channels based on the equivalent system, and determine a particular rate for the data transmission based on the metric.
 32. The receiver unit of claim 31, wherein the rate selector is further operative to estimate a spectral efficiency of each transmission channel based on the one or more estimated channel characteristics, and determine an average spectral efficiency of the transmission channels based on estimated spectral efficiencies of the transmission channels, and wherein the equivalent system is defined to have an additive white Gaussian noise (AWGN) channel and a spectral efficiency equal to the average spectral efficiency of the transmission channels.
 33. The receiver unit of claim 32, wherein the spectral efficiency of each transmission channel is estimated based on a constrained or unconstrained channel spectral efficiency function.
 34. The receiver unit of claim 32, further comprising: a memory configured to store one or more tables for a function used to estimate the spectral efficiency of each transmission channel.
 35. The receiver unit of claim 31, further comprising: a controller operative to provide feedback information comprised of the particular rate.
 36. An apparatus in a multi-channel communication system, comprising: means for identifying a plurality of transmission channels to be used for the data transmission; means for defining an equivalent system based on one or more estimated channel characteristics of the transmission channels; means for deriving a metric for the transmission channels based on the equivalent system; and means for determining a particular rate for the data transmission based on the metric.
 37. The receiver apparatus of claim 36, further comprising: means for estimating a spectral efficiency of each transmission channel based on the one or more estimated channel characteristics, and means for determining an average spectral efficiency of the transmission channels based on estimated spectral efficiencies of the transmission channels, and wherein the equivalent system is defined to have an additive white Gaussian noise (AWGN) channel and a spectral efficiency equal to the average spectral efficiency of the transmission channels.
 38. The receiver apparatus of claim 37, further comprising: means for storing one or more tables for a function used to estimate the spectral efficiency of each transmission.
 39. A transmitter unit in a multi-channel communication system, comprising: a controller operative to identify a rate to use for a data transmission over a plurality of transmission channels in a wireless communication channel, wherein the rate is determined based on an equivalent system defined for the transmission channels based on one or more estimated channel characteristics of the transmission channels; a transmit data processor operative to code data, provided at the identified rate, in accordance with a particular coding scheme to provide coded data; and a modulator operative to modulate the coded data in accordance with a particular modulation scheme to provide modulated data.
 40. The transmitter unit of claim 39, further comprising: a transmitter operative to generate at least one modulated signal for the modulated data.
 41. The transmitter unit of claim 39, wherein the multi-channel communication system is a MIMO communication system that employs OFDM, and the transmission channels correspond to frequency subchannels of spatial subchannels.
 42. An apparatus in a wireless communication system, comprising: means for identifying a rate to use for a data transmission over a plurality of transmission channels in a wireless communication channel, wherein the rate is determined based on an equivalent system defined for the transmission channels based on one or more estimated channel characteristics of the transmission channels; means for coding data, provided at the identified rate, in accordance with a particular coding scheme to provide coded data; and means for modulating the coded data in accordance with a particular modulation scheme to provide modulated data.
 43. A transmitter unit in a multi-channel communication system, comprising: a controller operative to identify a set of rates for a set of data streams to be transmitted over a wireless communication channel, wherein the rate for each data stream is determined based on an equivalent system defined for a group of transmission channels used for the data stream, and wherein the equivalent system for each transmission channel group is defined based on one or more estimated channel characteristics of the transmission channels in the group; at least one transmit data processor operative to code each data stream, provided at the identified rate, in accordance with a coding scheme selected for the data stream to provide a corresponding coded data stream; and at least one modulator operative to modulate each coded data steam in accordance with a modulation scheme selected for the data stream to provide a corresponding modulation stream.
 44. A multi-channel communication system comprising: a receiver unit including a channel estimator operative to derive estimates of one or more characteristics of a plurality of transmission channels, and a rate selector operative to define an equivalent system based on the one or more estimated channel characteristics of the transmission channels, derive a metric for the transmission channels based on the equivalent system, and determine a particular rate for the data transmission based on the metric; and a transmitter unit including at least one transmit data processor operative to code data, provided at the determined rate, in accordance with a coding scheme to provide coded data, and at least one modulator operative to modulate the coded data in accordance with a modulation scheme to provide modulated data. 